关于步进电机外文翻译

淮阴工学院

毕业设计(论文)外文资料翻译

系（院）：江淮学院

专业：电气工程及其自动化

姓名：朱佳

学号：3052110135

外文出处：Nonlinear Dynamics18:383-404,1999 (用外文写)

附件： 1.外文资料翻译译文；2.外文原文。

注：请将该封面与附件装订成册。

附件1：外文资料翻译译文

步进电机的振荡、不稳定以及控制

摘要：本文介绍了一种分析永磁步进电机不稳定性的新颖方法。结果表明，该种电机有两种类型的不稳定现象：中频振荡和高频不稳定性。非线性分叉理论是用来说明局部不稳定和中频振荡运动之间的关系。一种新型的分析介绍了被确定为高频不稳定性的同步损耗现象。在相间分界线和吸引子的概念被用于导出数量来评估高频不稳定性。通过使用这个数量就可以很容易地估计高频供应的稳定性。此外，还介绍了稳定性理论。广义的方法给出了基于反馈理论的稳定问题的分析。结果表明，中频稳定度和高频稳定度可以提高状态反馈。

关键词：步进电机，不稳定，非线性，状态反馈。

1. 介绍

步进电机是将数字脉冲输入转换为模拟角度输出的电磁增量运动装置。其内在的步进能力允许没有反馈的精确位置控制。也就是说，他们可以在开环模式下跟踪任何步阶位置，因此执行位置控制是不需要任何反馈的。步进电机提供比直流电机每单位更高的峰值扭矩;此外，它们是无电刷电机，因此需要较少的维护。所有这些特性使得步进电机在许多位置和速度控制系统的选择中非常具有吸引力，例如如在计算机硬盘驱动器和打印机，代理表，机器人中的应用等.

尽管步进电机有许多突出的特性，他们仍遭受振荡或不稳定现象。这种现象严重地限制其开环的动态性能和需要高速运作的适用领域。这种振荡通常在步进率低于1000脉冲/秒的时候发生，并已被确认为中频不稳定或局部不稳定[1]，或者动态不稳定[2]。此外，步进电机还有另一种不稳定现象，也就是在步进率较高时，即使负荷扭矩小于其牵出扭矩，电动机也常常不同步。该文中将这种现象确定为高频不稳定性，因为它以比在中频振荡现象中发生的频率更高的频率出现。高频不稳定性不像中频不稳定性那样被广泛接受，而且还没有一个方法来评估它。

中频振荡已经被广泛地认识了很长一段时间，但是，一个完整的了解还没有牢固确立。这可以归因于支配振荡现象的非线性是相当困难处理的。大多数研究人员在线性模型基础上分析它[1]。尽管在许多情况下，这种处理方法是有效的或有益

的，但为了更好地描述这一复杂的现象，在非线性理论基础上的处理方法也是需要的。例如，基于线性模型只能看到电动机在某些供应频率下转向局部不稳定，并不能使被观测的振荡现象更多深入。事实上，除非有人利用非线性理论，否则振荡不能评估。

因此，在非线性动力学上利用被发展的数学理论处理振荡或不稳定是很重要的。值得指出的是，Taft和Gauthier[3]，还有Taft和Harned[4]使用的诸如在振荡和不稳定现象的分析中的极限环和分界线之类的数学概念，并取得了关于所谓非同步现象的一些非常有启发性的见解。尽管如此，在这项研究中仍然缺乏一个全面的数学分析。本文一种新的数学分被开发了用于分析步进电机的振动和不稳定性。

本文的第一部分讨论了步进电机的稳定性分析。结果表明，中频振荡可定性为一种非线性系统的分叉现象（霍普夫分叉）。本文的贡献之一是将中频振荡与霍普夫分叉联系起来，从而霍普夫理论从理论上证明了振荡的存在性。高频不稳定性也被详细讨论了，并介绍了一种新型的量来评估高频稳定。这个量是很容易计算的，而且可以作为一种标准来预测高频不稳定性的发生。在一个真实电动机上的实验结果显示了该分析工具的有效性。

本文的第二部分通过反馈讨论了步进电机的稳定性控制。一些设计者已表明，通过调节供应频率[ 5 ]，中频不稳定性可以得到改善。特别是Pickup和Russell [ 6，7]都在频率调制的方法上提出了详细的分析。在他们的分析中，雅可比级数用于解决常微分方程和一组数值有待解决的非线性代数方程组。此外，他们的分析负责的是双相电动机，因此，他们的结论不能直接适用于我们需要考虑三相电动机的情况。在这里，我们提供一个没有必要处理任何复杂数学的更简洁的稳定步进电机的分析。在这种分析中，使用的是d-q模型的步进电机。由于双相电动机和三相电动机具有相同的d-q模型，因此，这种分析对双相电动机和三相电动机都有效。迄今为止，人们仅仅认识到用调制方法来抑制中频振荡。本文结果表明，该方法不仅对改善中频稳定性有效，而且对改善高频稳定性也有效。

2. 动态模型的步进电机

本文件中所考虑的步进电机由一个双相或三相绕组的跳动定子和永磁转子组成。一个极对三相电动机的简化原理如图1所示。步进电机通常是由被脉冲序列控制产生矩形波电压的电压源型逆变器供给的。这种电动机用本质上和同步电动机相同的原则进行作业。步进电机主要作业方式之一是保持提供电压的恒定以及脉冲频

率在非常广泛的范围上变化。在这样的操作条件下，振动和不稳定的问题通常会出现。

图1.三相电动机的图解模型

用q–d框架参考转换建立了一个三相步进电机的数学模型。下面给出了三相绕组电压方程

v a = Ri

a

+ L*di

a

/dt ? M*di

b

/dt ? M*di

c

/dt + dλ

pma

/dt ,

v b = Ri

b

+ L*di

b

/dt ? M*di

a

/dt ? M*di

c

/dt + dλ

pmb

/dt ,

v c = Ri

c

+ L*di

c

/dt ? M*di

a

/dt ? M*di

b

/dt + dλ

pmc

/dt , (1)

其中R和L分别是相绕组的电阻和感应线圈，并且M是相绕组之间的互感线圈。

λpma, λpmb and λpmc是应归于永磁体的相的磁通，且可以假定为转子位置的正弦函数如下

λpma = λ1 sin(Nθ),

λpmb = λ1 sin(Nθ? 2/3),

λpmc = λ1 sin(Nθ - 2/3), (2) 其中N是转子齿数。本文中强调的非线性由上述方程所代表，即磁通是转子位置的非线性函数。

使用Q ,d转换，将参考框架由固定相轴变换成随转子移动的轴（参见图2）。矩阵从a,b,c框架转换成q,d框架变换被给出了[8]

(3)

例如，给出了q,d参考里的电压

(4)

在a,b,c参考中，只有两个变量是独立的（i

a + i

b

+ i

c

= 0），因此，上面提到

的由三个变量转化为两个变量是允许的。在电压方程（1）中应用上述转换，在q,d

框架中获得转换后的电压方程为

v q = Ri

q

+ L

1

*di

q

/dt + NL

1

i

d

ω + Nλ

1

ω,

v d = Ri

d

+ L

1

*di

d

/dt ?NL

1

i

q

ω, (5)

图2，a,b,c和d,q参考框架

其中L1 = L + M，且ω是电动机的速度。有证据表明,电动机的扭矩有以下公式

T = 3/2Nλ

1i

q

. (6)

转子电动机的方程为

J*dω/dt = 3/2*Nλ

1i

q

?B

f

ω–Tl , (7)

如果B

f

是粘性摩擦系数，和Tl代表负荷扭矩（在本文中假定为恒定）。

为了构成完整的电动机的状态方程，我们需要另一种代表转子位置的状态变量。为此，通常使用满足下列方程的所谓的负荷角δ[8]

Dδ/dt = ω?ω

, (8)

其中ω

是电动机的稳态转速。方程（5），（7），和（8）构成电动机的状态空间

模型，其输入变量是电压v

q 和v

d

.如前所述，步进电机由逆变器供给，其输出电压不

是正弦电波而是方波。然而，由于相比正弦情况下非正弦电压不能很大程度地改变振荡特性和不稳定性（如将在第3部分显示的，振荡是由于电动机的非线性），为了本文的目的我们可以假设供给电压是正弦波。根据这一假设，我们可以得到如下的

v q 和v

d

v

q

= V

m

cos(Nδ) ,

v

d

= V

m

sin(Nδ) , (9)

其中V

m

是正弦波的最大值。上述方程，我们已经将输入电压由时间函数转变为

状态函数，并且以这种方式我们可以用自控系统描绘出电动机的动态，如下所示。这将有助于简化数学分析。

根据方程（5），（7），和（8），电动机的状态空间模型可以如下写成矩阵式

? = F(X,u) = AX + Fn(X) + Bu , (10) 其中X = [iq id ω δ] T , u = [ω1 Tl] T 定义为输入，且ω1 = N ω0 是

供应频率。输入矩阵B 被定义为

矩阵A 是F(.)的线性部分，如下

F n (X)代表了F(.)的线性部分，如下

输入端u 独立于时间，因此，方程（10）是独立的。

在F(X,u)中有三个参数，它们是供应频率ω1，电源电压幅度V m 和负荷扭矩T l 。

这些参数影响步进电机的运行情况。在实践中，通常用这样一种方式来驱动步进电机，即用因指令脉冲而变化的供应频率ω1来控制电动机的速度，而电源电压保持不

变。因此，我们应研究参数ω1的影响。

3.分叉和中频振荡，

设ω=ω0，得出方程（10）的平衡

且φ是它的相角，

φ = arctan(ω1L1/R) . (16) 方程（12）和（13）显示存在着多重均衡，这意味着这些平衡永远不能全局稳

定。人们可以看到，如方程（12）和（13）所示有两组平衡。第一组由方程（12）

对应电动机的实际运行情况来代表。第二组由方程（13）总是不稳定且不涉及到实

际运作情况来代表。在下面，我们将集中精力在由方程（12）代表的平衡上。

附件2：外文原文

Oscillation, Instability and Control of Stepper Motors

LIYU CAO and HOWARD M. SCHWARTZ

Department of Systems and Computer Engineering, Carleton University, 1125 Colonel By Drive,

Ottawa, ON K1S 5B6, Canada

(Received: 18 February 1998; accepted: 1 December 1998) Abstract. A novel approach to analyzing instability in permanent-magnet stepper motors is presented. It is shown that there are two kinds of unstable phenomena in this kind ofmotor: mid-frequency oscillation and

high-frequency instability. Nonlinear bifurcation theory is used to illustrate the relationship between local instability and midfrequency oscillatory motion. A novel analysis is presented to analyze the loss of synchronism phenomenon, which is identified as high-frequency instability. The concepts of separatrices and attractors in phase-space are used to derive a quantity to evaluate the high-frequency instability. By using this quantity one can easily estimate the stability for high supply frequencies. Furthermore, a stabilization method is presented. A generalized approach to analyze the stabilization problem based on feedback theory is given. It is shown that the mid-frequency stability

and the high-frequency stability can be improved by state feedback. Keywords: Stepper motors, instability, nonlinearity, state feedback.

1. Introduction

Stepper motors are electromagnetic incremental-motion devices which convert digital pulse inputs to analog angle outputs. Their inherent stepping ability allows for accurate position control without feedback. That is, they can track any step position in open-loop mode, consequently no

feedback is needed to implement position control. Stepper motors deliver higher peak torque per unit weight than DC motors; in addition, they are brushless machines and therefore require less maintenance. All of these properties have made stepper motors a very attractive selection in many position and speed control systems, such as in computer hard disk drivers and printers, XY-tables, robot manipulators, etc.

Although stepper motors have many salient properties, they suffer from an oscillation or unstable phenomenon. This phenomenon severely restricts their open-loop dynamic performance and applicable area where high speed operation is needed. The oscillation usually occurs at stepping rates lower than 1000 pulse/s, and has been recognized as a mid-frequency instability or local instability [1], or a dynamic instability [2]. In addition, there is another kind of unstable phenomenon in stepper motors, that is, the motors usually lose synchronism at higher stepping rates, even though load torque is less than their pull-out torque. This phenomenon is identified as

high-frequency instability in this paper, because it appears at much higher frequencies than the frequencies at which the mid-frequency oscillation occurs. The high-frequency instability has not been recognized as widely as mid-frequency instability, and there is not yet a method to evaluate it.

Mid-frequency oscillation has been recognized widely for a very long time, however, a complete understanding of it has not been well established. This can be attributed to the nonlinearity that dominates the oscillation phenomenon and is quite difficult to deal with.

384 L. Cao and H. M. Schwartz

Most researchers have analyzed it based on a linearized model [1]. Although in many cases, this kind of treatments is valid or useful, a treatment based on nonlinear theory is needed in order to give a better description on this complex phenomenon. For example, based on a linearized model one can only see that the motors turn to be locally unstable at some supply frequencies, which does not give much insight into the observed oscillatory

phenomenon. In fact, the oscillation cannot be assessed unless one uses nonlinear theory.

Therefore, it is significant to use developed mathematical theory on nonlinear dynamics to handle the oscillation or instability. It is worth noting that Taft and Gauthier [3], and Taft and Harned [4] used mathematical concepts such as limit cycles and separatrices in the analysis of oscillatory and unstable phenomena, and obtained some very instructive insights into the socalled loss of synchronous phenomenon. Nevertheless, there is still a lack of a comprehensive mathematical analysis in this kind of studies. In this paper a novel mathematical analysis is developed to analyze the oscillations and instability in stepper motors.

The first part of this paper discusses the stability analysis of stepper motors. It is shown that the mid-frequency oscillation can be characterized as a bifurcation phenomenon (Hopf bifurcation) of nonlinear systems. One of contributions of this paper is to relate the midfrequency oscillation to Hopf bifurcation, thereby, the existence of the oscillation is proved theoretically by Hopf theory. High-frequency instability is also discussed in detail, and a novel quantity is introduced to evaluate high-frequency stability. This quantity is very easy

to calculate, and can be used as a criteria to predict the onset of the high-frequency instability. Experimental results on a real motor show the efficiency of this analytical tool.

The second part of this paper discusses stabilizing control of stepper motors through feedback. Several authors have shown that by modulating the supply frequency [5], the midfrequency

instability can be improved. In particular, Pickup and Russell [6, 7] have presented a detailed analysis on the frequency modulation method. In their analysis, Jacobi series was used to solve a ordinary differential equation, and a set of nonlinear algebraic equations had to be solved numerically. In addition, their analysis is undertaken for a two-phase motor, and

therefore, their conclusions cannot applied directly to our situation, where a three-phase motor will be considered. Here, we give a more elegant analysis for stabilizing stepper motors, where no complex mathematical manipulation is needed. In this analysis, a d–q model of stepper motors is used. Because two-phase motors and three-phase motors have the same q–d model and therefore, the analysis is valid for both two-phase and three-phase motors. Up to date, it is only recognized that the modulation method is needed to suppress the midfrequency oscillation. In this paper, it is shown that this method is not only valid to improve mid-frequency stability, but also effective to improve high-frequency stability.

2. Dynamic Model of Stepper Motors

The stepper motor considered in this paper consists of a salient stator with two-phase or threephase windings, and a permanent-magnet rotor. A simplified schematic of a three-phase motor with one pole-pair is shown in Figure 1. The stepper motor is usually fed by a voltage-source inverter, which is controlled by a sequence of pulses and produces square-wave voltages. This

motor operates essentially on the same principle as that of synchronous motors. One of major operating manner for stepper motors is that supplying voltage is kept constant and frequency

of pulses is changed at a very wide range. Under this operating condition, oscillation and instability problems usually arise.

Figure 1. Schematic model of a three-phase stepper motor.

A mathematical model for a three-phase stepper motor is established using q–d framereference transformation. The voltage equations for three-phase windings are given by

v a = Ri

a

+ L*di

a

/dt ? M*di

b

/dt ? M*di

c

/dt + dλ

pma

/dt ,

v b = Ri

b

+ L*di

b

/dt ? M*di

a

/dt ? M*di

c

/dt + dλ

pmb

/dt ,

v c = Ri

c

+ L*di

c

/dt ? M*di

a

/dt ? M*di

b

/dt + dλ

pmc

/dt ,

where R and L are the resistance and inductance of the phase windings, and M is the mutual inductance between the phase windings. _pm a, _pm b and _pm c are the flux-linkages of the

phases due to the permanent magnet, and can be assumed to be sinusoid functions of rotor position _ as follow

λpma = λ1 sin(Nθ),

λpmb = λ1 sin(Nθ? 2/3),

λpmc = λ1 sin(Nθ - 2/3),

where N is number of rotor teeth. The nonlinearity emphasized in this paper is represented by the above equations, that is, the flux-linkages are nonlinear functions of the rotor position.

By using the q; d transformation, the frame of reference is changed from the fixed phase axes to the axes moving with the rotor (refer to Figure 2). Transformation matrix from the a; b; c frame to the q; d frame is given by [8]

For example, voltages in the q; d reference are given by

In the a; b; c reference, only two variables are independent (ia C ib C ic D 0); therefore, the above transformation from three variables to two variables is allowable. Applying the above

transformation to the voltage equations (1), the transferred voltage equation in the q; d frame can be obtained as

v q = Ri

q

+ L

1

*di

q

/dt + NL

1

i

d

ω + Nλ

1

ω,

v d =Ri

d

+ L

1

*di

d

/dt ? NL

1

i

q

ω, (5)

Figure 2. a, b, c and d, q reference frame.

where L1 D L C M, and ! is the speed of the rotor.It can be shown that the motor’s torque has the following form [2]

T = 3/2Nλ

1i q

The equation of motion of the rotor is written as

J*dω/dt = 3/2*Nλ

1i

q

? B

f

ω– Tl ,

where Bf is the coefficient of viscous friction, and Tl represents load torque, which is assumed to be a constant in this paper.

In order to constitute the complete state equation of the motor, we need another state variable that represents the position of the rotor. For this purpose the so called load angle _ [8] is usually used, which satisfies the following equation

Dδ/dt = ω?ω

,

where !0 is steady-state speed of the motor. Equations (5), (7), and (8) constitute the statespace model of the motor, for which the input variables are the voltages vq and vd. As mentioned before, stepper motors are fed by an inverter, whose output voltages are not sinusoidal but instead are square waves. However, because the non-sinusoidal voltages do not change the oscillation feature and instability very much if compared to the sinusoidal case (as will be shown in Section 3, the oscillation is due to the

nonlinearity of the motor), for the purposes of this paper we can assume the supply voltages are sinusoidal. Under this assumption, we can get vq and vd as follows

v q = V

m

cos(Nδ) ,

v d = V

m

sin(Nδ) ,

where Vm is the maximum of the sine wave. With the above equation, we have changed the input voltages from a function of time to a function of state, and in this way we can represent the dynamics of the motor by a autonomous system, as shown below. This will simplify the mathematical analysis.

From Equations (5), (7), and (8), the state-space model of the motor can be written in a matrix form as follows

? = F(X,u) = AX + Fn(X) + Bu , (10) where X D T iq id ! _U T , u D T!1 Tl U T is defined as the input, and !1 D N!0 is the supply frequency. The input matrix B is defined by

The matrix A is the linear part of F._/, and is given by

Fn.X/ represents the nonlinear part of F._/, and is given by

The input term u is independent of time, and therefore Equation (10) is autonomous.

There are three parameters in F.X;u/, they are the supply frequency !1, the supply voltage magnitude Vm and the load torque Tl . These parameters

govern the behaviour of the stepper motor. In practice, stepper motors are usually driven in such a way that the supply frequency !1 is changed by the command pulse to control the motor’s speed, while the supply voltage is kept constant. Therefore, we shall investigate the effect of parameter !1.

3. Bifurcation and Mid-Frequency Oscillation

By setting ! D !0, the equilibria of Equation (10) are given as

and ' is its phase angle defined by

φ = arctan(ω1L1/R) . (16) Equations (12) and (13) indicate that multiple equilibria exist, which means that these equilibria can never be globally stable. One can see that there are two groups of equilibria as shown in Equations (12) and (13). The first group represented by Equation (12) corresponds to the real operating conditions of the motor. The second group represented by Equation (13) is always unstable and does not relate to the real operating conditions. In the following, we will concentrate on the equilibria represented by Equation (12).