Anti-windup_schemes_for_proportional_integral_and_proportional_resonant_controller
九年级地理环境保护英语阅读理解20题
九年级地理环境保护英语阅读理解20题1<背景文章>Global warming is one of the most serious environmental issues facing our planet. It refers to the long-term increase in the average temperature of the Earth's climate system. The phenomenon of global warming is caused by a variety of factors. One of the main causes is the increase in greenhouse gas emissions, such as carbon dioxide, methane, and nitrous oxide. These gases trap heat in the atmosphere and cause the Earth's temperature to rise.The effects of global warming are far-reaching. Rising sea levels threaten coastal areas and low-lying islands. More frequent and intense extreme weather events, such as hurricanes, floods, and droughts, can cause significant damage to human lives and property. Changes in precipitation patterns can also affect agriculture and water resources.To address global warming, we need to take action on multiple fronts. We can reduce greenhouse gas emissions by using renewable energy sources, such as solar and wind power. We can also improve energy efficiency in buildings, transportation, and industry. In addition, we can protect and restore forests, which absorb carbon dioxide from the atmosphere.1. What is global warming?A. A short-term increase in temperature.B. A long-term increase in the average temperature of the Earth's climate system.C. A decrease in temperature.D. No change in temperature.答案：B。
PID Control System Analysis and Design
➢ Increases the phase-lag
➢ Gain margin (GM) and phase margin (PM) are reduced, and the closed-loop system becomes more oscillatory and potentially unstable
Integrator Windup Remedies
➢ Antiwindup can be achieved implicitly through automatic reset.
➢ Explicit Antiwindup implemented explicitly through internal negative feedback.
STANDARD STRUCTURES OF PID CONTROLLERS
➢ A PID controller is a phase lead-lag compensator with one pole at the origin and the other at infinity.
➢ PI-Phase lag.
➢ Unstable system
Remedies
➢ Involves use of filters
Linear low pass filter Velocity Feedback SetPoint Filter Nonlinear median filter
Linear low pass filter
➢ Windup is due to the controller states becoming inconsistent with the saturated control signal, and future correction is ignored until the actuator desaturates
anti-Windup Compensator Designs for Nonsalient Permanent-Magnet Synchronous Motor Speed Regulators
Anti-Windup Compensator Designs for Nonsalient Permanent-Magnet Synchronous MotorSpeed RegulatorsPhil March,Member,IEEE,and Matthew C.TurnerAbstract—This paper presents and compares a number of anti-windup compensator designs which address the problem of current saturation within high-performance permanent-magnet synchronous motor applications.The compensator variants in-clude an integrator reset scheme,back calculation and tracking, and an optimally synthesized low-order dynamic compensator. Performance is comparedfirst through simulation tests using a nonlinear nonsalient machine model with multirate discrete time control loops.Second,the results of tests repeated on a hardware implementation are shown.In both simulation and experiment,the back calculation and tracking and low-order designs are shown to exhibit clear performance advantages over the reset strategy.Index Terms—Anti-windup(A W),constraints,current limita-tion,nonsalient,optimal,permanent-magnet synchronous motor (PMSM),saturation,speed regulation.N OMENCLATUREi a,i b,i c Stator phase currents.i d Stator current vector d-axis component.i q Stator current vector q-axis component.V a,V b,V c Stator phase voltages.V d Stator voltage vector d-axis component.V q Stator voltage vector q-axis component.θe Rotor electrical position.θm Rotor mechanical position.ωm Rotor speed(mechanical).ψf Flux linkage.load Rotor load torque.T Electromagnetic torque.P Number of magnetic pole pairs.φPhase advance angle.K e Motor back-EMF constant.B Damping coefficient.R s Stator phase winding resistance.L s Stator phase winding inductance.J Rotor moment of inertia.Paper2008-IDC-078.R1,presented at the2007IEEE International Electric Machines and Drives Conference,Antalya,Turkey,May3–5,and approved for publication in the IEEE T RANSACTIONS ON I NDUSTRY A PPLICATIONS by the Industrial Drives Committee of the IEEE Industry Applications Society. Manuscript submitted for review September18,2008and released for publica-tion March8,2009.Current version published September18,2009.This work was supported in part by the U.K.Engineering and Physical Sciences Research Council and in part by TRW Automotive.The authors are with the Department of Engineering,University of Leicester, Leicester,LE17RH,U.K.(e-mail:phil.march@;mct6@). Color versions of one or more of thefigures in this paper are available online at .Digital Object Identifier10.1109/TIA.2009.2027157I.I NTRODUCTIONW ITHIN permanent-magnet synchronous motor(PMSM) control systems,certain constraints are required to safe-guard the motor and the power electronics;for instance,restric-tions on the magnitude of voltages and currents.The inclusion of these constraints in the control system introduces saturation nonlinearities to the plant inputs and states.These saturation constraints can cause controller“windup”problems when the controller attempts to exceed these during operation and lead to performance degradation,particularly in high-performance applications where saturation is more prevalent.Therefore,for maximum performance to be obtained,there is the requirement to consider these saturation constraints in the design of PMSM control systems.The term windup was originally conceived to describe the unrestricted accumulation of an integral controller,observed when a plant input limit renders a set-point reference infea-sible.Since then,the term has been used more broadly,and is used to refer to more general performance degradation ef-fects associated with the violation of plant input limits.This larger class of undesirable saturation effects includes not only the aforementioned“integrator windup”but also includes the performance degradation resulting from saturation with modern control designs that do not have an explicit integral state,and performance degradation with coupled multivariable systems where saturation of one input can affect a number of perfor-mance outputs.A number of different methods,which vary significantly in complexity of design and implementation,could be applied to this problem.Perhaps the simplest approach is to use a low-gain linear controller,which will lead to saturation being encountered less frequently.However,due to linearity,this is achieved typically at the expense of limiting performance.An alternative way of avoiding saturation would be to use H∞techniques[1],[2]to produce(possibly multivariable)controllers which can distrib-ute control effort in a more efficient manner.The drawback to both of the aforementioned approaches is that,since they are linear,they treat large-and small-signal behavior in exactly the same way and thus cannot be used to achieve high-performance small-signal behavior,and large-signal behavior which is less prone to saturation simultaneously.An alternative to the above is to use model-predictive-control[3],[4](MPC)techniques in which the control constraints are incorporated into an online finite horizon optimization procedure,producing controllers which respect the saturation constraints directly.Of course the0093-9994/$26.00©2009IEEEproblem with H∞and,particularly,MPC techniques is that they tend to require more online computation than standard classical control schemes and thus are not well suited to cost sensitive applications.The approach we consider in this paper is referred to as anti-windup(AW)compensation and corresponds to the sec-ond stage in a two-stage controller design process.In stage one,a“nominal”controller is designed to meet the desired performance objectives in the absence of saturation.This con-troller governs the closed loop dynamics exclusively while the saturation constraint is respected.In stage two,a so-called AW compensator is designed with which the control system is augmented.This compensator only becomes active during saturation,and is designed to maintain stability of the closed loop system and minimize the degradation of performance when violation of the saturation nonlinearity leads to the pres-ence of complex nonlinear dynamics.Models of the constraints are included in the controller and saturation of these software constraints activates the compensator to deal with the saturation event.This approach allows for the nominal controller to be designed to provide good small signal performance,and the AW compensator designed subsequently to handle saturation and therefore ensure good large-signal behavior also.Another benefit of AW is that many compensator designs can commonly be implemented as static feedback gains or dynamicfilters with just a few states,and therefore do not make large computational demands on the control computer.Thus,in many ways,AW may be the most logical and appealing solution.A large number of AW schemes are in existence and the reader is referred to[5]–[10]and references therein for an overview of some well-known approaches.One crucial advantage of AW is that it only modifies the existing controller’s behavior during and immediately following saturation events and so does not require an existing controller to be completely redesigned.II.S YSTEM U NDER C ONSIDERATIONIn this paper,we apply AW to deal with violation of satura-tion constraints in the model of a PMSM speed control system for an automotive electric actuator.A.PMSM ModelThe PMSM used is a surface mounted design and therefore has very low saliency.Thus,its dynamics can be reproduced quite accurately by the nonsalient model of(1)–(4)in the d–q coordinate system[11]di d dt =1L s[V d−R s i d+P L sωm i q](1)di q dt =1L sV q−R s i q−P L sωm i d−K eωm√3(2)dωm dt =1J[T−load−Bωm](3)T=√32K e i q.(4)A description of the symbols used for this model and for an equivalent model in three phase coordinates[11],[12]is found in the nomenclature.AW design is normally based on linear models of the plant and controller under consideration,and thus we desire to extract a linear model of the plant.To this end,the aforementioned model is linearized about an equilibrium(trim) pointωm(0),i d(0),i q(0),with corresponding input signals V d(0),V q(0),and load(0),giving the linear state-space model of(5)and(6)in deviation variables.It is important to note that the dynamics of the linear model are dependent upon the trim conditions and are most sensitive to changes in the trim speed.Thus,the speed at which the model is trimmed is an important consideration for developing an appropriate linear model.It should also be noted that,as the motor has several inputs,there is not a unique trim point at a given speed;a wide range(a continuum)of voltage vectors can achieve the same trim speed.In addition,the applied load also affects the speed achieved and therefore the trim point.For this reason, trim points were determined from closed-loop simulations with the given controller in place and were chosen to coincide with the steady-state values enforced by this controllerddt⎡⎣δi dδi qδωm⎤⎦=⎡⎢⎣−R sL sPωm(0)P i q(0)−Pωm(0)−R sL sP i d(0)−K eL s√30K e√32J−BJ⎤⎥⎦×⎡⎣δi dδi qδωm⎤⎦+⎡⎣01/L s0001/L s−1/J00⎤⎦⎡⎣δloadδV dδV q⎤⎦(5)δωm=[001]⎡⎣δi dδi qδωm⎤⎦.(6)B.Controller StructureThe constrained three-phase motor control system which we desire to apply AW compensation to is shown in Fig.1. Clarke and Park transformations[12]are used to convert the three-phase current measurements into d–q-axis coordinates, allowing control to be computed in the rotor reference frame, prior to conversion back into three-phase signals to be applied to the plant.The control system consists of an inner current control loop and an outer speed control loop.The inner loop contains two independent PI regulators,used to control the magnitude of the stator current components in the direct and quadrature axes by manipulating the stator voltage vector in d–q coordinates.The outer loop controller is shown in Fig.2and generates the d-and q-axis current demands that are passed to the inner loop. For maximum torque operation,the d-axis current demand is set to zero and a PI regulator in the q-axis exploits the linear relationship between electromagnetic torque and q-axis current(4)to provide speed(ωm)tracking capabilities by manipulating the q-axis current demand.This provides the most efficient operation and enables speeds up to base speed to be attained.For operation beyond base speed,the nonlinear d-axis portion of the speed controller comes into effect which ad-vances the phase of the current demand vector byφelectrical degrees from the q-axis.The angleφis predetermined as a nonlinear function of speed and is usually read from a lookupid _dmd 2+iq _dmd 2≤i _lim.(7)To convert this “multivariable”type of saturation into a more standard saturation problem,it is possible to consider separate scalar saturation functions on the d -and q -axis currentFig.4.d,q-axis current limit.ing knowledge of the phase advance angleφas a function of speed,these limits can be varied separately to restrict the magnitude of the current demand vector idq_dmd to the limit i_lim,while maintaining the desired phase.In this case,we have−i d≤id_dmd≤i d(8)−i q≤iq_dmd≤i q(9) and the d-and q-axis limits¯i d and¯i q are determined by trigonometry to be sin(φ)i_lim and cos(φ)i_lim,respectively. Note that the current limit has now been simplified from satu-ration of the norm of a vector signal to saturation of two scalar signals,albeit with time varying limits.A graphical depiction of how these d-and q-axis limits vary with the phase advance angle in rotor electrical coordinates is shown in Fig.4.A useful property of this representation of norm saturation is that violation of the circular limit implies that both the d-and q-axis components of the vector exceed their own limits. Conversely,if the q-axis limit is not exceeded,nor is the d-axis limit,and the vector demand lies within the circle.This link between d-and q-axis saturation can be exploited,allowing AW to be applied only to the q-axis.This follows since only the q-axis portion of the speed controller is dynamic,having no states which contribute to windup.The compensator is designed to manipulate the speed controller states during and immediately following saturation,leading the q-axis output, and hence also the d-axis output,out of saturation.This means that for AW design,only the q-axis scalar saturation function need be considered.The d-axis saturation function can then be incorporated as an additional nonlinearity into the d-axis portion of the speed controller.Note that during operation at speeds below0.22normalized units,the d-axis controller is not active and henceφ=0.In this case,the current limit is purely enforced on the q-axis current.III.AW C OMPENSATIONA block diagram showing how a generic AW compensation scheme is applied to the speed control loop is shown in Fig.5.A signal˜u is defined as the difference between saturated and unsaturated control signals,indicating the existence and severity of a saturation event when˜u=0,and normal operation when˜u=0.This signal drives the compensator block AW in some manner to modify the action of the linear q-axis portion of the controller and solve the windup problem.Three AW com-pensation schemes are introduced in this section which interact with the linear portion of the controller in different ways.As previously mentioned,for the work here,it is possible simply to consider the q-axis current saturation for AW design. Furthermore,we have also neglected the nonlinear d-axis con-troller in the design of the AW compensator.Part of the reason for this was that the d-axis controller is not actually active until a speed of0.22normalized units is attained and thus for a por-tion of the motor’s operational envelope is effectively absent. Second,as the d-axis controller is,by its very nature,nonlinear, it is more difficult to handle this within the conventional AW framework.Finally,the linear q-axis controller is arguably the dominant part of the controller and extensive simulation revealed that it was sufficient to consider this component alone for AW design.The authors do accept that the neglect of the d-axis portion of the controller does yield a small gap between the theory of AW compensator design and its application,but this did not appear to be detrimental in either the simulation or test results.To put the design and function of the three compensator candidates in context,wefirst introduce a simulation for which saturation without AW causes problematic performance.For the existing control system,a large step change in speed causes saturation of the current limit,as shown in the simulation results of Fig.6.This particular demand causes the q-axis control signal to exceed the limit for the majority of the simulation, leading to poor tracking of the high-speed demand and an overshoot of the subsequent low-speed demand.Note that the q-axis current limit varies with speed,and that only the signals within the controller exceed the limit since the signals passed onto the current controller are limited.A.Reset AWA simple way of preventing integrator windup in PI con-trollers is to reset the integrator state when saturation is ex-perienced and therefore to prevent the integrator accumulating excessive“energy.”When saturation ceases,integration is then allowed to continue either from the previously held value or another value such as zero.For the method considered in this paper,the integrator state is recalculated during saturation such that the controller output is held at the saturation level.When saturation ceases,integration recommences from the value calculated at the previous sample instance and normal linear operation continues.This is described by(10)and(11)where iq_lim represents the q-axis current limit,kp and ki are the speed controller proportional and integral gains,respectively, I(k)and e(k)denote the integrator state and speed error at sample instance k,respectively,andτspd represents the sample period of the speed controlleriq_dmd(k)=kp e(k)+I(k)(10) I(k)=⎧⎨⎩iq_lim−kp e(k),˜u>0I(k−1)+kiτspd e(k),˜u=0−iq_lim−kp e(k),˜u<0.(11)Fig.6.Saturating system without AW.Methods such as this are commonly used in industry to prevent integrator windup,and this method or variations on the theme can be found preprogrammed in off-the-shelf control hardware.This form of AW can be appealing since it is simple to implement and requires no tuning,although its performance could be improved upon.B.Back Calculation and Tracking(BCAT)In this method,the AW compensator adopts the form of a scalar feedback gain F,which is introduced between the signal˜u and the input to the integrator in the PI controller (Fig.7).This feedback reduces the integrator state over a period of time while saturation continues and will therefore prevent integrator windup.The magnitude of the gain F determines the rate at which the integrator state is reduced and can be tuned to improve performance.This method,and systematic ways for choosing F,is described more fully in[13].The main benefit of this method is its simple transparent architecture and simple tuning.It can be applied successfully in many applications, although it is not well suited to multivariable systems and stability guarantees are not easily acquired.It may also not be directly applicable to controllers without an explicit integral state,for example,many H∞designs.Note that the nonlinear d-axis portion of the speed controller has been omitted from the block diagram of Fig.7for simplicity.C.Low-Order Dynamic AWThis method differs from the others in that it is model based, and as such provides the potential for improved performance by accounting for the dynamics of the plant and controller in the design of the compensator.This compensator adopts the form of two discretefiltersΘ1andΘ2which modify the control action at both the input and output of the nominal controller,as shown in Fig.8.Although thesefilters are discrete,we choose to useand in particular,it was shown that for low-order synthesis M(s)needed to be chosen asM=(I−K2G2)−1(−K2Θ2+Θ1+I)whereΘ1(s)Θ2(s)=F1(s)˜Θ1F2(s)˜Θ2and F1(s),F2(s)are some appropriate transfer functions cho-sen by the designer and˜Θ1,˜Θ2are gain matrices which are synthesized in an optimal fashion.Moreover,in[15],a linear-matrix-inequality(LMI)method was proposed which allowed˜Θ1,˜Θ2to be synthesized such that an upper bound on theL2gain of the operator T p:u lin→y d was minimized.This procedure allows the AW compensator,shown in Fig.8,to be synthesized in an optimal manner.The results presented in this paper are produced using a low-order compensator which has been synthesized in continuous time and then discretized. However,discrete-time synthesis routines are also in existence, as shown in[16].Unlike the other methods hitherto described,low-order AW provides stability and performance guarantees for the closed-loop system and also is applicable to multiple-input–multiple-output systems,for which simpler strategies often perform poorly.The disadvantage of this method is that,given arbitrary filters F1(s),F2(s),no a priori guarantees of the existence of a compensator are given;this must be checked using the LMI-optmization procedure.However,previous work has indicated thatfirst-orderfilters often prove to be effective choices,and once a successful set offilters have been found,the bandwidths can be varied by the designer to tune for the best overall performance.If a feasible solution cannot be found for the low-order synthesis problem,it is always possible tofind a globally stabilizing AW compensator of order equal to the plant using the methods presented in[17]provided that the plant is asymptotically stable.IV.S IMULATION-B ASED A NALYSISFor simulation tests,the following PMSM model in the stator reference frame[11],[12]is used:di a dt =1L sV a−R s i a+K e√3ωm sin(θe)(12)di b dt =1L sV b−R s i b+K e√3ωm sinθe−2π3(13)di c dt =1L sV c−R s i c+K e√3ωm sinθe−4π3(14)dθedt=Pωm(15)dωm dt =1J[T e−load−Bωm](16)T=−K e√3i a sinθ+i b sinθ−2π3+i c sinθ−4π3.(17)The performance of the three AW designs were comparedusing a simulation model built using the aforementioned motormodel,inverse Park and Clarke transformations[12],and thediscrete time control structure presented in this paper.The innercurrent control loop is run at a fast sample rate in order tofaithfully reproduce the sinusoidal phase voltages which havea frequency of P times the maximum motor speed.However,there are still small errors in the stator phase voltages due tosampling and this does affect performance.This is particularlytrue at high speed when the frequency of the sine wave to beapplied becomes closer to the controller sampling frequencysince the errors become larger.With the d–q axis model,thehigh-frequency sinusoids are not reproduced so these samplerate issues are not revealed.A.AW ActionFigs.10–12show the output and control responses of thespeed control system with each type of AW to the same speeddemand as used for Fig.6.With the reset scheme,the q-axis demand exceeds the saturation limit but the integrator isreset at the next sample instance,successfully bringing thecontrol signal to below the limit(Fig.10).However,becausethe proportional action of the controller diminishes as themotor accelerates,the controller output then drops below thesaturation level for a time,leading to a slower rise time.Asimilar effect is observed during the reverse step.While thisresponse is significantly better than without AW(Fig.6),it isnot ideal.This undesirable feature of reset AW is dependentupon the system dynamics and the tune of the PI controller andcan usually be eliminated by reducing the proportional gain.However,this restricts the tuning of the controller and mayimpair the small signal response of the system.With the back calculation and tracking method,the integratoris reset gradually,as observed by the smooth decay of the q-axis demand down to the saturation level for both forward andreverse steps(Fig.11).This leads to longer periods of saturationthan with the reset design,but performance is significantly bet-ter overall,showing that bringing the system out of saturationas quickly as possible may not be the best approach.The low-order compensator functions differently in that thecompensator can directly alter the controller output.This meansthat bringing the control signal down to the saturation level isless important in the prevention of windup.Note that while thecontrol signal may greatly exceed the saturation limit(Fig.12),the system gracefully recovers tracking performance when sat-uration ceases.B.Simulation Performance ComparisonsResponses to a doublet speed demand of the system aug-mented by each form of AW are shown in Fig.13.The ben-efit of AW compensation can clearly be seen;performance isimproved by all compensator types,although the reset schemeis less effective at limiting performance degradation.Both the“back calculation and tracking”and the“low order”designsachieve significantly faster rise times for both positive and neg-ative step speed demands.They also perform very similarly onFig.10.Saturating system with resetAW. Fig.11.Saturating system with BCAT AW.the simulation model,despite their differing design processes and implementation.When a constant load torque is applied such that the full speed reference is infeasible,the designs perform as shown in Fig.14.For the system without AW,the infeasible reference causes an error signal to persist and the integrator continues to accumulate,leading to a q-axis current demand far in excess of the saturation level.When the reverse step is demanded,the integrator state holds the control signal in saturation for a period of time,resulting in a significant delay in the response.Again, performance is improved by the application of each form of AW,with the back calculation design performing best,closelyFig.12.Saturating system with low-orderAW.Fig.13.AW responses to a doublet speed demand under no load conditions. followed by the low-order design,and both outperforming the reset strategy.V.E XPERIMENTAL A NALYSISFor practical testing of the AW designs,the controller code was written using TargetLink rapid prototyping software and flashed to a16-bitfixed-point arithmetic processor using Vector CANape.A sophisticated pulsewidth modulation(PWM)algo-rithm was adopted to generate the stator phase voltages.This incorporates the use of third harmonic injection techniques to maximize voltageutilization.Fig.14.AW responses to a doublet speed demand under static loading.A.Model AccuracyIt is inevitable that there will be differences between the performance of the model and that of the real system due to parametric uncertainty and unmodeled dynamics.An example of such a discrepancy is shown in Fig.15where performance of the practical system and the simulation model with reset AW are compared in response to the speed reference of Fig.6.In the practical system,the decelerating step reference is rate limited to prevent the motor operating as a generator.This prevents a deceleration demand from causing saturation,and so the adverse effects observed in the simulation response areFig.15.Experimental and simulated performance comparison under no load. not seen in practice.This focuses the analysis of experimental AW performance on speed references which cause the motor to accelerate.For the accelerating step reference,the experimental data show a faster response with a small amount of overshoot. The main source of this discrepancy is considered to be the modeling of the hydraulic load for which a simple linear approximation is used.In practice,this is highly nonlinear and difficult to model accurately.In addition,the PWM strategy is modeled simply by inverse Park and Clarke transformations and a simple gain adjustment.This simplification is expected to be least accurate when maximum voltage is demanded,for instance in response to large demands.Another implication of modeling errors relates to the com-pensator design stage.Since the low-order compensator is model based,any significant inaccuracies in the model could cause the design to perform more poorly than predicted when applied to the real system.However,the low-order design is observed to perform well in both cases.In addition,the performance trends observed during simulation tuning were also evident in the experimental tests,i.e.,the best performing design on the simulation model also performed best on the real system,giving confidence that the fundamental dynamics of the model are correct.B.Implementation IssuesThe use offixed-point arithmetic requires afixed range and scaling to be assigned for each signal,state,and gain in the controller.Inevitably,errors are introduced to the representation of each number,but these can become significant when the number to be represented is small compared to the range.These errors can propagate through the controller and accumulate over time due to successive arithmetic operations,and so to minimize their effect,consideration needs to be given to the scaling assigned to each signal/state/gain.This is particularly true when implementing state-spacefilters as used in low-order dynamic compensation.A givenfilter can be represented by an arbitrary number of state-space realizations with the same input output relationship,but different state and output equation matrices.Infixed point,the choice of state-space realization can affect the accuracy of computations frominput Fig.16.Experimental data—large step speed demand under static load.to state and from state to output,and hence the accuracy of the input output relationship can vary.A further consideration is that errors in the state to output computation will only be present for one sample instance,but errors in the input to state computation will also affect future sample instances since the error is retained in the state.These factors must be carefully considered when selecting the state-space representation to be used,but a good starting point is a balanced realization.For the low-order compensator we present,thefilters chosen are stable,strictly proper,and offirst-order low-pass design.These characteristics make the choice of adequate scalings relatively simple.C.Experimental Performance ComparisonsIn this section,we compare the performance of the three AW designs as observed using the experimental rig.Current demand saturation is induced for quite small step speed demands while operating at high speed,and when large step demands are made at lower speeds.Unless particular comment is made,the performance differences are very similar under no load and static load conditions.Fig.16shows experimental performance comparisons be-tween the different AW controllers for a large step demand under a small static load.While the reset scheme reduces the overshoot compared to the case without AW,performance is improved further by the BCAT method,and further still by the low-order dynamic design.Certainly,overshoot is caused by the baseline controller,but this is accentuated by saturation,and this component can be improved by good AW design.Fig.17shows step tracking performance for a small step demand in the high-speed range.In this case,the reset and BCAT strategies provide very similar performance but the low-order dynamic compensator performs best,roughly halving the overshoot compared with the previous two designs.VI.C ONCLUSIONThe performance of“back calculation and tracking”and “low-order dynamic”AW compensation has been compared to a reset scheme commonly used in industry through simu-lation tests and a practical implementation.Both compensation。
小学上册第十三次英语第3单元测验试卷
小学上册英语第3单元测验试卷英语试题一、综合题(本题有100小题，每小题1分，共100分.每小题不选、错误，均不给分)1.The _____ (绿色能源) movement promotes plant-based solutions.2.The _____ (cat/dog) is sleeping.3.The ______ (香味) of flowers can be very strong.4.What do we call a scientist who studies the properties of matter?A. ChemistB. PhysicistC. BiologistD. Geologist答案:A5.I want to _____ (see/watch) a movie tonight.6.What is the main language spoken in the USA?A. SpanishB. FrenchC. EnglishD. German答案:C7.The cake is ________ and sweet.8.We share ______ (秘密) with each other.9.I enjoy going ______ (远足) to enjoy the beauty of nature.10.The ______ teaches us about civic responsibilities.11.The chemical properties of an element depend on its ______ structure.12.The country famous for its tango is ________ (阿根廷).13.What is the opposite of 'fast'?A. QuickB. SlowC. SpeedyD. Rapid14.What is the capital of Tunisia?A. TunisB. SousseC. MonastirD. Bizerte答案:A15.What do you call a house made of ice?A. IglooB. CabinC. CastleD. Hut答案:A16._____ (当地植物) can adapt to specific environments.17.Chemical reactions often involve a change in ______.18.The __________ is the layer of skin that helps to protect against injury.19.I enjoy making memories with my toy ________ (玩具名称).20.I can ________ (manage) my time effectively.21.What do we call the opposite of ‘clean’?A. DirtyB. NeatC. TidyD. Clear22. A __________ is a sudden release of energy in the Earth's crust.23.My friend is a ______. He loves to collect stamps.24.My dad shows me how to be __________ (勇敢的) in difficult times.25.What is the largest country in the world?A. CanadaB. ChinaC. RussiaD. India26.The _______ (The Battle of Waterloo) marked the defeat of Napoleon.27.The ____ is a small creature that hides under leaves.28.What do we call the study of the universe?A. BiologyB. AstronomyC. GeologyD. Physics答案:B29.The _____ (果实) develops after a flower blooms.30. Wall of China was built to __________ (保护) the country from invasions. The Grea31.The ant carries food back to its ______ (巢).32.What is 4 + 5?A. 8B. 9C. 10D. 1133.The _____ (水果收成) happens in late summer.34.Many plants have a specific ______ period for blooming. (许多植物有特定的开花期。
INTERNATIONAL JOURNAL OF CIRCUIT THEORY AND APPLICATIONS
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Figure 1. Block diagram of a PFC boost pre-regulator.
Copyright ? 2003 John Wiley & Sons, Ltd. Int. J. Circ. Theor. Appl. 2003; 31:611–624
SUMMARY This paper investigates the fast-scale instability in a power-factor-correction (PFC) boost converter under a conventional average current-mode control. The converter is operated in continuous mode. Computer simulations and theoretical analysis are performed to study the e ects of the time-varying input voltage under the variation of some chosen parameters on the qualitative behaviour of the system. It is found that fast-scale instability may occur during a line cycle, which can cause distortion to the line current and degrade the practical power factor. The results provide useful information for the design of PFC boost converters to avoid distortion due to fast-scale bifurcation. Copyright ? 2003 John Wiley & Sons, Ltd.
高一经济现象英语阅读理解30题
高一经济现象英语阅读理解30题1<背景文章>The market economy is an economic system in which decisions regarding production, distribution, and consumption are guided by the interactions of supply and demand. In a market economy, businesses and individuals are free to make their own economic decisions.One of the key characteristics of a market economy is the role of competition. Competition among businesses leads to lower prices, better quality products, and greater efficiency. When businesses compete, they are forced to find ways to produce goods and services more efficiently in order to lower costs and offer better prices to consumers.Another important aspect of a market economy is the price mechanism. Prices play a crucial role in allocating resources. When the demand for a particular good or service increases, its price tends to rise. This signals to producers that there is a greater need for that product, and they respond by increasing production. Conversely, when the demand for a product falls, its price drops, and producers reduce production.The market economy also promotes innovation. Businesses are constantly looking for new ways to improve their products and processes in order to gain a competitive edge. This leads to the development of newtechnologies and better ways of doing things.In addition, a market economy allows for a wide range of choices for consumers. With many businesses competing for their business, consumers have the opportunity to choose from a variety of products and services at different prices and quality levels.1. One of the key characteristics of a market economy is _______.A. government controlB. lack of competitionC. the role of competitionD. fixed prices答案：C。
NOVATEK-ELECTRO LTD 产品说明书：UBZ-301（10-100A）通用感应电机保护
NOVATEK-ELECTRO LTDResearch-and-Manufacture CompanyUBZ-301 (10-100A)UNIVERSAL INDUCTIONMOTOR PROTECTION UNITUSERS MANUALCONTROLS DESCRIPTION AND DIMENSIONS DIAGRAM 4151. Control for nominal current setting, Inom;2. Control for operating current setting, Iop (% Inom);3. Control for T2 (double overload trip) delay setting;4. Combined trip adjustment control for Umin/Umax;5. Control for phase imbalance adjustment, PI;6. Control for trip threshold for the minimum current, Imin (%Inom);7. Control for automatic reset delay setting, Ton;8. Green LED indicating for the mains voltage presence;9,10,11. Red LEDs indicating faults for insulation, overload and U fault respectively;12. Green LED indicating for load energization;13. Output terminals;14. Input terminals (10,11,12 are used for the connection with the BO-01 exchange unit);15. Insulation monitoring terminal.1 APPLICATIONSThe UBZ-301(10-100A)universal induction motor protection unit is designed for the continuous monitoring of the mains voltage parameters and for RMS phase/line currents of 3-Phase AC 380V/50Hz electrical equipment monitoring, primarily, of induction motors whose power is from 5kW up to 50 kW, isolated neutral system included.The unit provides full and effective protection of electrical equipment by a magnetic starter (contactor) coil control.The unit isolates electrical equipment from the running system and/or disables its start. This is performed in the following cases:1. when the mains voltage is of poor quality (unallowable voltage jumps, phase loss, incorrect phase sequence and phase «coincidence», phase/line voltage imbalance);2. when mechanical overloads (symmetrical phase/line current overload) take place. The unit performs overload protection with a dependent time delay;3. when phase/line current asymmetrical overloads induced by faults inside the motor occur. The unit performs protection from phase current imbalance and further disables an automatic reset;4. when phase current asymmetry without overload occurs that is induced by the insulation fault inside the motor and/or the power cable;5.when motor load is lost(«dry stroke»for pumps).The unit provides the minimum start and/or operating current protection;6. when insulation level to frame is abnormally low. The unit performs insulation level test before start and if the insulation is poor the start is disabled.7.when stator winding ground-to-fault occurs during operation. The unit performs the ground leakage current protection.The UBZ-301 (10-100A) provides:•a simple and accurate electromotor nominal current setting by nominal current standard scale;•the electromotor operating current setting that differs from standard values;•overload tripping with a dependent time delay (the current-time characteristic curve is plotted for a conventionally cold motor). The motor heat balance differential equation is being solved in the operation process. This approach enables to take account of the preceding electromotor status and to make a decision on heat overload presence with the maximum validity. This method also permits to allow for a motor start heating and to restrict (at the customer’s option) amount of starts per unit time;•shift of current-time characteristic curve along the current-axis and along time-axis as well;•setting of trip thresholds for the minimum/the maximum voltage,line voltage&phase current imbalance, and also for automatic reset delay at the personal customer’s discretion;•fault type indication, the mains voltage presence indication, current range indication the unit is adjusted to, and load energization indication;•the data exchange and transfer to the local computer network according to the RS-485 MODBUS record through the BO-01 exchange unit (BO-01 is supplied on order).2 DESCRIPTIONThe unit is a microprocessor-based digital device that provides a high degree of reliability and accuracy. The unit doesn’t need any auxiliary supply because it retrieves it's energy demand out of the measurement signal: it’s self-powered by the voltage to be monitored. Simultaneous isolated independent monitoring for the mains voltage and phase currents permits to detect the type of occurring fault and to provide a different decision-making logic for each fault type. When the mains voltage faults occur the unit performs automatic load reset on return voltage parameters to normal operating conditions. If a fault is induced by abnormal condition inside the motor(phase current imbalance at the symmetrical mains voltage,leakage current presence etc.) restart is disabled.The unit is stocked with three toroidal current transducers. Two of them are the phase/line current transducers (TT1, TT2), power phase cables are pulled through them. The third transducer is the differential current transducer(DCT)that has an enlarged diameter,because three power phase cables are pulled through it. By the 6, 7, 8, 9 terminals the unit is connected in parallel to the mains supply to be monitored. The unit output is provided with N.O. and N. C. contacts (the 1, 2, 3, 4 terminals). The output 3-4 terminals are connected in series with the starter coil power supply (with control circuit). The5terminal is designed to monitor the insulation level. The unit wiring diagram is shown below.When the unit trips the load is de-energized by a break in the magnetic starter coil power circuit through the N. C. 3-4 contacts.Table 1 - The 1-2-3-4 output contacts specificationMax. current for~ 250 V A. C.Max. powerMax sustained safevoltage ~Max. current for U = 30V D.C.Cosφ = 0.43A 2000VA 460V 3ACosφ = 1.05ANominal parameters and trip thresholds are set by front-panel screwdriver potentiometers.Nominal current setting. Nominal current is set by № 1 potentiometer. There are eleven positions of the potentiometer. Each position corresponds to the specific standard nominal current scale value (see below Table of Nominal Currents). Each position is characterized by the specific number of blinks that the green «On» LED makes. To set the nominal current one needs to bring out potentiometer control arm to a corresponding position; when the unit is energized the number of blinks «On» LED must correspond to the Table below. One needs to take into account that there are «dead bands» between the positions where «On» LED glows without blinks and where the nominal current is indefinite.In order to set operating value which is different from the nominal one that is specified in the nominal current table, it’s recommended the № 1 potentiometer to set to the position corresponding to the nearest value from the nominal current scale, and by the № 2 potentiometer one can increase or decrease the necessary value in % from the set value.Table 2 - Nominal current tablePotentiometer №1 devisionsNom. current, АGreen LED «On» blink1101bl.- pause 212,52bl.- pause 3163bl.- pause 4204bl.- pause 5255bl.- pause 6326bl.- pause 7407bl.- pause 8508bl.- pause 9639bl.- pause 108010bl.- pause 1110011bl.- pauseNOTE S:1.Continuous green «On» LED glow means that the potentiometer is set in «dead band». One needs to set the potentiometer so as the green LED blinked and the number of blinks corresponded to the set nominal current.2.Nominal currents setting is to be performed correcting for load connections (Wye/Delta), according to ratings of engine.Controls and adjustmentsThe unit has seven independent controls. For user’s convenience screwdriver slots of adjusting potentiometers are brought out to the unit front panel.•№1 – «Inom» - nominal current setting; there are eleven positions and each position corresponds to the specific current from the nominal currents table;•№2 – «Iop» - operating current; it is set in ± 15 percent of nominal current, it has ten scale marks;•№3 – «T2» - overload trip delay when there is double overload for operating current set; in the central position T2 ≈ 58-60 seconds The minimum time delay is 10 seconds, the maximum time delay is 100 seconds. The control shifts current-time characteristic dependence along time axis;•№4 – «Unom(%)» - combined control for Umax/Umin threshold in percent of the nominal voltage; according to this setting before the load energization the unit is checking the mains voltage level and, depending on its value, permits or forbids the load energization; after the load has been energized the voltage monitoring is going on but the load de-energization decision is made for currents;•№5 – «PI%» - trip threshold control for line voltage imbalance and RMS phase current imbalance; it has ten scale marks. The parameter is calculated as the difference between the maximum and the minimum values, in percent of the maximum value. If current imbalance percentage is twice as much as voltage imbalance percentage then it’s supposed that the imbalance is induced by fault conditions inside the motor. The automatic reset is forbidden and the unit is disabled;•№6 – «Imin» - trip threshold control for the minimum operating current, in percent of operating current set. It has ten scale marks from 0% to 75%: in «0» position this control is off;•№7 – «Ton» - automatic reset delay, it is within 0 – 600 seconds range; the scale is logarithmic.Indication•the green «ON» LED indicates that voltage exists in the mains. In the blink mode of glow the blink number between pauses corresponds to the specific nominal current from the nominal current table; there is a continuous glow in a «dead band». One needs to set a nominal current in the blink mode of operation;•the green «Load» LED indicates that the load is energized (the 3-4 terminals are closed);•the red «Insulation» LED lights up with continuous glow before the start if the stator/ power cable winding insulation level is abnormally low (less than 500 kOhms), and also during operation when there is a tripping for differential current; the unit is disabled;•the red «U Fault» LED glows when the mains voltage fault has occurred. The blink mode of operation switches on when there is undervoltage/overvoltage, phase imbalance for the mains voltage, voltage is not present on all three phases;•incorrect phase sequence or phase coincidence induces the mode of operation when all three red LEDs are blinking in turn;•the red «Overload» LED blinks when the average phase current exceeds the nominal one. After the unit has tripped for overload this LED comes to glow during 0.9 AR (automatic reset) delay.4 TECHNICAL BRIEFNominal line voltage, V380Mains frequency, Hz45-55Nominal current range in UBZ-301 10-100, А10-100Operating current setting range, % of nominal±15Double overload delay adjustment range, sec10-100Voltage threshold adjustment range, % of nominal±(5-20)Phase imbalance adjustment range, %5-20Trip threshold adjustment range for the minimum current, % of nominal0-75Automatic reset delay adjustment range ( Тon), sec0-600First energization load delay when Тon= 0, sec2-3Trip delay for current overload According to current-time characteristic curveTrip delay for voltage fault, sec2Trip delay for current fault (overload excluded), sec2 Fixed trip point for leakage current, А 1.0 Insulation resistance threshold, kОhms500±20 Voltage hysteresis, V10/17 Heat hysteresis, % of stored-up heat after load de-energization33Trip threshold accuracy for current, % of nominal current, not more than2-3 Trip threshold accuracy for voltage, V, not more than3 Phase imbalance accuracy, %, not more than 1.5 Operating voltage range, % of nominal one50-150 Power consumption (under load), VA, not more than 3.0 Maximum switched current of output contacts, A5 Output contact life:- under 5A load , operations, no less than - under 1A load , operations, no less than 100 000 1 000 000Enclosure:- apparatus- terminal block IP40 IP20Operating temperature range, °C from -35 to +55 Storage temperature, °C from -45 to +70 Weight, kg, not more than0.200Case dimensions 4 modules of S-typeMounting standard 35 mm DIN-railMounting position arbitrary5 OPERATION1. After supply voltage has been applied to the unit and before the output relay is energized the unit checks:•a stator winding insulation level to frame. If insulation resistance is below 500±20 kOhms, the load is not energized. The red «Insulation» LED glows;•the mains voltage quality,i.e.if voltage is present on all three phases,if the mains voltage is symmetrical, what the RMS line voltage value is like. When any of inhibit factors is present, the load is not energized, the red «U Fault» LED blinks;•a correct phase sequence, and phase «non-coincidence». When any of inhibit factors is present, the load is not energized, all red LEDs are blinking in turn; If all the parameters are normal, the outlet relay will be energized after Ton delay has expired (the 3-4 contacts are being closed and the 1-2 contacts are being opened) - the green «Load» LED glows. If load currents are absent (there are no less 2% of nominal one) the reason is that the load is de-energized. Voltage monitoring and insulation level is going. Output relay of unit is de-energized if inhibit factors are present in pause without currents;2. After the load is energized the unit performs voltage and current monitoring. The decision on load de-energization is made according to the following factors:•RMS current exceeds the nominal (operating) current (set by №№ 1, 2, 3 potentiometers); if there is current overload without heat overload the red «Overload» LED blinks but the load is not de-energized. If current overload induces heat overload the load is de-energized. The red «Overload»LED glows and is ON during 0.9Ton. The automatic reset is permitted;•current imbalance (set by №5 potentiometer) is twice exceeds the mains voltage imbalance; the load is de-energized, all red LEDs glow, the unit is disabled, the automatic reset is forbidden. To enable the unit one needs to remove supply voltage from the unit. It’s supposed that this type of fault is induced by abnormal conditions inside the motor;•current imbalance (set by №5 potentiometer) is less than twice exceeds voltage imbalance; the load is de-energized, the red «U Fault» LED glows, the automatic reset is permitted;•current imbalance (set by №5 potentiometer) is less than voltage imbalance; the load is de-energized, the red «U Fault» LED blinks, the automatic reset is permitted;•the average current value is less than Imin (set by №6 potentiometer); the load is de-energized, all red LEDs blink simultaneously, the unit is disabled, the automatic reset is forbidden. To enable the unit one needs to remove the supply voltage from the unit.Electromotor protection against heat overloadThe electromotor heat balance equation is being solved as the work advances. It’s supposed that:•the motor was cold before start;•during operation the motor releases the heat which is proportional to the current square;•after the stop the motor cools down exponentially.Below is the current-time characteristic curve with different T2 values (set by №3 potentiometer), where:•I/In – current ratio relative to the nominal current;•T/T2 -- actual trip delay relative to T2 (set by № 3 potentiometer).Current-time characteristic dependenceThe current-time characteristic dependence shown in the tables below is given for the standard recommended T2 value (the №3 potentiometer middle position corresponds to 60 seconds when double overload occurs):I/Inom 1.1 1.2 1.4 1.72 2.73456781015Тsec36524714888.66036.424.613.58.5 5.9 4.3 3.3 2.10.9After the load has been de-energized owing to the heat overload it will automatically be energized again:•according to heat hysteresis if time delay Ton=0, i. e. the motor must cool down 33% of the stored up heat;•according time delay Ton (№ 7 potentiometer) if Ton isn’t equal 0.By suitable selection of different Ton values, heat hysteresis considered, one can reduce number of starts per time unit because in the intermittent cycle the unit stores heat quantity released at the start of the motor.6 PRELIMINARY STARTING PROCEDURE AND SERVICE MANUALThe unit produced is completely ready for operation and needs no special pre-starting procedure measures. Owing to digital technology all the unit settings are aligned quite accurate, so no control devices are needed to adjust them. Use of the unit according to specifications above and the present service manual, continuous work included, relieves of preventive maintenance during service life. To put the unit in operation one must follow operating instructions given below:1.To set nominal (operating) current, trip thresholds, trip delays and reset delay by potentiometer's contact arms.2.To connect the unit according to the wire diagram given below:•by the 6(L1), 7(L2), 8(L3), 9(N) terminals the unit is connected in parallel to the mains supply to be monitored;•two current transducers (each one of them is put on two power phase wires that carry the load) are connected to the 13, 14, 15, 16 terminals; in connecting one has to consider the transducers grading;1st transducer– the beginning – the 13 terminal, the end – the 14 terminal;2nd transducer– the beginning – the 15 terminal, the end – the 15 terminal;•a differential current transducer that is put on the three power phase wires must be connected to the 17, 18 terminals (the connection grading is unimportant);•the 5 insulation monitoring terminal is connected to one of the MS output contacts;•output contacts (the 3-4 terminals) are connected to the MS coil power supply circuit (control circuit);•the BO-01 exhange and date transfer unit is connected to the 10, 11, 12 terminals (this unit is supplied on order).3.To apply a voltage to the unit. The correct setting of nominal current is checked by the number of blinks that the green LED makes. After Ton has expired (if there are no factors that can forbid energizing) the output relay of the unit is energized. If Ton=0, the first energizing will occur after 2-3 sec delay has expired.NOTE - The unit must be connected subject to the safety regulations. To set settings is recommended in «off» state. To set settings alive is permitted in the test conditions subject to the safety regulations.ATTENTION!If immediately after the load has been energized the unit de-energizes it and disables it for current imbalance,the incorrect polarity of the current transducers TT1or TT2 connection may be a reason. Then it’s recommended to change one of the transducers connection by reversing the places“the beginning-the end”of the13-16terminals.If the effect pointed above repeats when the load is re-energized it means that the transducers were connected correctly and the imbalance arose from EM and/or power cable fault.NOTES:1 Transducers are mounted by plastic clamps (they are component parts of supplies).2 The phase wires which are passing through the differential transducer to try to symmetrize in the centerof the transducer.WIRING DIAGRAMDT–differential current transducer (differential current transformer);CT1,CT2–current--transducers;BO-01– exchange and date transfer unit (on order)NOTE:1 “START”-button and “STOP”-button can be connected to MSC power supply circuit if necessary;2 The 220V MSC connection is shown here. The 380V MSC power supply circuit is analogous, coil power is applied from different phases through the 3-4 contacts;3 If BO-01 is absent the 10, 11, 12 terminals are not used.7 STORAGE AND SHIPPING CONDITIONSThe unit in manufacturer package should be stored in enclosed rooms at –45 to +70 °C and exposed to no more than 80% of relative humidity when there are no fumes in the air that exert a deleterious effect on package and the unit material. The Buyer must provide the protection of the unit against mechanical damages in transit.8 WARRANTYNovatek-Electro LTD. Company warrants a trouble-free operation of the UBZ-301 (10-100A) unit manufactured by it within 36 months from the date of sale, provided:•the proper installation;•the safety of the inspection quality control department seal;•the integrity of the case, no traces of an opening, cracks, spalls etc.。
2023年12月英语六级听力原文及参考答案
2023年12月英语六级听力原文及参考答案听力稿原文section AConversation 1气候变化和全球经济发展W: Professor Henderson could you give us a brief overview of what you do, where you work and your main area of research？M: Well the Center for Climate Research where I work links the science of climate change to issues around economics and policy。
Some of our research is to do with the likely impacts of climate change and all of the associated risks。
W: And how strong is the evidence that climate change is happening that it‘s really something we need to be worried about。
M: Well most of the science of climate change particularly that to do with global warming is simply fact。
But other aspects of the science are less certain or at least more disputed。
And so we‘re really talking about risk what the economics tells us is thatit’s probably cheaper to avoid climate change to avoid the risk than it has to deal with the likely consequences。
2025年北师大版高考英语试题与参考答案
2025年北师大版英语高考复习试题与参考答案一、听力第一节（本大题有5小题，每小题1.5分，共7.5分）1、Listen to the following dialogue between two students, and answer the question.Student A: Hey, are you planning to follow the exam schedule strictly? Student B: Yeah, I always try to stick to a routine. How about you?Student A: Well, I like to mix it up a bit. It keeps me motivated.Question: What does Student A prefer when it comes to following an exam schedule?A. To follow the routine strictly.B. To mix up the schedule to stay motivated.C. To follow the schedule only when it’s convenient.D. To avoid any schedule altogether.Answer: BExplanation: Student A indicates that they like to mix up the schedule to stay motivated, which is equivalent to choice B.2、 Listen to the following conversation about a school trip, and complete the following sentence with the correct information.Teacher: Ok, everyone, we’re going to have a field trip next week. It’s a science-themed trip to the museum downtown.Student A: That sounds amazing! What are we going to learn there, though?Teacher: Well, you’ll get a behind-the-scenes look at how exhibits are put together, and you’ll interact with some of the curators. Plus, there are interactive displays where you can try out different experiments.Question: What will the students be able to do during the trip to the museum?A. Simply observe the exhibits without participating.B. Work with the curators to put together new exhibits.C. Participate in interactive experiments and discussions.D. Finish the field trip without visiting the museum.Answer: CExplanation: The teacher mentions that the students will be able to participate in interactive experiments and discussions, which corresponds to choice C.3.What does the man suggest doing?A) Having a picnic.B) Going to the cinema.C) Visiting the museum.D) Playing tennis.Answer: A) Having a picnic.Explanation: The woman mentions that it’s a beautiful day and asks the man what he thinks they should do. The man responds by suggesting they take advantage of the weather and have a picnic in the park. Therefore, the correct answer isA) Having a picnic.4.Where are the speakers most likely?A) At home.B) In a restaurant.C) On a bus.D) In a bookstore.Answer: B) In a restaurant.Explanation: The dialogue involves one speaker asking for recommendations on dishes and commenting on the menu, while the other speaker provides suggestions and describes the specials. This context strongly suggests that the conversation is taking place in a restaurant, making B) In a restaurant the correct choice.5、 Listening Section AQuestion: How is the woman going to the airport?A) By bus.B) By taxi.C) By subway.Answer: BExplanation:In the recording, the man asks, “Are you going to the airport by bus or by taxi?” The woman replies, “I decide to take a taxi because it will be faster.” Therefore, the correct answer is B) By taxi.解析：录音中，男士问：“你要去机场是乘公交还是打车？”女士回答：“我决定打车去，因为会更快。
IEC61400-1-2005风电机组设计要求标准英汉对照
需要什么文档直接在我的文档里搜索比直接在网站大海捞针要容易的多也准确省时的多
INTERNATIONAL STANrbines – Part 1:
Design requirements
Publication numbering As from 1 January 1997 all IEC publications are issued with a designation in the 60000 series. For example, IEC 34-1 is now referred to as IEC 60034-1.
Further information on IEC publications The technical content of IEC publications is kept under constant review by the IEC, thus ensuring that the content reflects current technology. Information relating to this publication, including its validity, is available in the IEC Catalogue of publications (see below) in addition to new editions, amendments and corrigenda. Information on the subjects under consideration and work in progress undertaken by the technical committee which has prepared this publication, as well as the list of publications issued,is also available from the following: IEC Web Site (www.iec.ch) Catalogue of IEC publications The on-line catalogue on the IEC web site (www.iec.ch/searchpub) enables you to search by a variety of criteria including text searches,technical committees and date of publication. Online information is also available on recently issued publications, withdrawn and replaced publications, as well as corrigenda. IEC Just Published This summary of recently issued publications (www.iec.ch/online_news/justpub) is also available by email. Please contact the Customer Service Centre (see below) for further information. Customer Service Centre If you have any questions regarding this publication or need further assistance, please contact the Customer Service Centre: Email: custserv@iec.ch Tel: +41 22 919 02 11 Fax: +41 22 919 03 00 .
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Anti-windup Schemes for Proportional Integral and Proportional Resonant ControllerAnirban Ghoshal and Vinod JohnDepartment of Electrical Engineering,Indian Institute of Science,BangaloreEmail:anirban@ee.iisc.ernet.in,vjohn@ee.iisc.ernet.inAbstract—Problems like windup or rollover arise in a PI controller working under saturation.Hence anti-windup schemes are necessary to minimize performance degradation.Similar situation may occur in a Proportional Resonant(PR)controller in the presence of a sustained error input.Several methods can be employed based on existing knowledge on PI controller to counter this problem.In this paper few such schemes are proposed and implemented in FPGA and MATLAB and from the obtained results their possible use and limitations have been studied.Index Terms—Anti-windup,Proportional resonant con-trol,PI control,Current control.I.I NTRODUCTIONIn a PI controller windup or rollover is a widely studied problem[1–5].Typically this problem arises if the input error to the controller is large or the input error remains nonzero for a long time.The controller output may saturate either because of large error input or because of nonzero error which makes the integrator output keep on accumulating.A controller under sat-uration may give delayed response to any change in the input and this delay would be more if the con-troller goes into deeper saturation level.Again digital implementation of controller using DSP or FPGA having fixed word length asks for careful choice of per unit values and saturation level of the controller to prevent windup phenomena.Hence it becomes necessary to employ anti-windup strategy to prevent the controller from going into deep saturation and to check windup or rollover of controller output.Anti-windup schemes for PI controller are a well researched topic though newer and innovative methods are still coming up.Several such schemes for PID controllers utilizing limited or conditional integration method and tracking anti-windup method have been discussed in[1].A modified tracking anti-windup method for PI controllers used in variable speed motor drives is described in[2].Design of an anti-windup scheme utilizing the benefits of both conditional integration and tracking or back-calculation method is shown in[3].A survey on current controlled techniques [4]shows that for control of electric drives and grid connected inverter systems,popular controller strategies use PI controllers.Anti-windup strategies for PI con-trollers used in such applications were under study for long.In recent times current control strategies such as Proportional Resonant(PR)Controller are gaining popu-larity for their advantages over PI controller particularly for control of grid connected inverters[6,7].Here the controller operates on AC signal whereas in popular d-q control or the synchronous reference frame based control the PI controller works on DC signal.It can be shown that under certain condition a PR controller can become prone to windup phenomena.In this paper studies on few‘Anti-windup’schemes for PI controller and their applicability in solving windup problem in PR controller are discussed.These methods were simulated in MATLAB and then implemented on a FPGA based digital controller.This digital controller board uses Al-tera’s‘Cyclone’series FPGA.A word length of16bit, 1pu=3F F F h which is equivalent to5V when viewed through DAC and sampling rate of10kHz were used for implementation.II.PI CONTROLLER WITH ANTI-WINDUPA.PI controllerTransfer function of a PI controller is expressed asG P I(s)=K p+K iFig. 1.Block diagram representation of a PI controller with saturation.So the variables can attain values between certain finite range and any value out of that range will lead to roll over and output of controller (y )would show it as a sudden jump from positive to negative or vice versa.Such undesirable condition can arise if the error (e )is too large or it remains non-zero for long duration during which the integrator causes the roll over.To limit the output a saturation block can be used at the outputterminal as shown in Fig.1.Here output (y ′)can be expressed asy ′=yfor Y max >y >Y min Ymax for y ≥Y max Yminfor y ≤Y min(3)But the above approach would lead to problem if theinput ‘e ’remains non-zero for long and during that period the integrator output keeps on accumulating.This may either introduce delay in response when the input error changes or even lead to roll over of output ‘y ’as can be seen in Fig.2.To avoid this it is necessaryFig.2.Output of PI controller with saturation implemented in FPGA for error input e =1.25.Here Kp =1.33and Ki =20.7.Upper saturation limit Y max =5V and lower saturation limit Y min =−5V .to check the integration process during such situations which is in general known as anti-windup.In this paper study has been carried out on two schemes using condi-tional integration and tracking anti-windup.Discussionson these commonly adopted anti-windup schemes for PI controllers are in the next subsections.B.Conditional IntegrationThe first method,shown in Fig.3is by stopping the integration process when the output y has reached the saturation limit.This method can also be considered asFig.3.Block diagram representation of a PI controller with anti-windup scheme described in Conditional Integration.a variable structure PID control method discussed in [5].This ensures that while the controller is experiencing saturation there is no further increase in the value of output ‘y ’.If the error reduces below certain level for which output comes out of saturation,the integrator starts working again.The results are shown in Fig.4.In this method the PI controller output never goes beyond the saturation limit.Upon reaching saturation the value at which the integrator output is held,depends on input er-ror magnitude and the proportional constant.A variation of this method can be by forcing the integrator output to some predetermined value whenever the controller output getssaturated.Fig. 4.Output of PI controller with anti-windup scheme of Conditional Integration implemented in FPGA for an error input e =1.25.Here Kp =1.33and Ki =20.7.Upper saturation limit Y max =5V and lower saturation limit Y min =−5V .C.Tracking Anti-windupFig.5shows another way of controlling the extent of saturation.Here the difference of actual output (y )andsaturated output (y ′)is fed back through a gain (K lim )to reduce the amount of error input e going intointegrator.Fig.5.Block diagram representation of a PI controller where thedifference between actual output (y )and saturated output (y ′)is used as a feedback signal through a gain (K lim )to reduced the input to integrator.Thus rate at which the output of integrator would have increased is reduced.Under normal situation the anti-windup path does not contribute in any way as y =y ′.The results are shown in Fig.6.Choice of limiting gain (K lim )depends on acceptable restriction on integrator output.Higher value of K lim keeps the actual output (y )close to the saturated output (y ′)which in turn enables the controller to come out of saturation quickly when theFig.6.Output of PI controller with anti-windup scheme of Tracking Anti-windup implemented in FPGA for an error input e =1.25.Here K p =1.33,K i =20.7and K lim =1.Upper saturation limit Y max =5V and lower saturation limit Y min =−5V .The steady state value of output ‘y ’is of interest for choosing K lim .Considering a situation where y >Y max following relationships can be derived to get an indi-cation of steady state and dynamic behaviour of the PI controller under saturation with tracking anti-windupcontrol.If the output of integrator is x ,thendxdt=−K i K lim x +K i (1−K lim K p )e +K i K lim Y max(6)Solution to the above differential equation for a constant error input e (t )=E gives x (t )=(X 0−EKlim+Y max −K p ×E )(7)Here X 0is the initial value of state ‘x ’.Now output ‘y ’can be expressed as y (t )=(X 0−EKlim+Y max )(8)If steady state value of ‘y ’is Y ss ,thenY ss =Y max +Es 2+ω2(10)HeresK iFig.7.Frequency response of the resonant part of a PR controller.Here K i =2000and ω=314rad/sec .(13)for a stable realisation are obtained using Eulers modified method.Here ∆T is the integration step size.p [n +1]=p [n ]+K i ×∆T ×e [n ]+ω×∆T ×q [n ](11)q [n +1]=q [n ]−ω×∆T ×p [n +1](12)y [n +1]=K p ×e [n ]+p [n +1](13)Fig.8.Block diagram representation of PR controller.Present literatures on PR controller shows several other ways of realisation.In the most commonly used form the resonant part is realized by a second order transfer function with a non-zero damping factor.In this realisation the resonant controller output is bounded for any sinusoidal input but error input is never actually zero.Here choice of damping factor would decide on error magnitude.Also,design of controller should take account of maximum allowable error input for a given damping factor to prevent roll over.The way of imple-mentation where damping factor is zero,shown in Fig.8and implemented by equations (11),(12)and (13),works fine as the error ‘e ’goes to zero at steady state.Butif error ‘e ’,typically a sinusoidal signal,persists for long because of some operating situations then output of resonant part can show unstable behaviour.The time domain expression for output of resonant part i.e.‘p ’,can be expressed as follows p (t )=A 0sin(ωt +φ0)+A ×K i ×t2ωsin(ωt )sin(φ)(14)where,e (t )=A ×sin(ωt −φ)(15)A 0=Q 0)(17)P 0and Q 0are the initial conditions of the states p and q .The above expression shows that if the error is present continuously then output of resonant part would keep on increasing and after sometime the entire controller output y would experience roll over.So it becomes necessary for the controller to limit the output under such situations.Therefore use of anti-windup logic need to be looked into.In this regard the schemes that find use in case of a PI controller can be used with a PR controller and their suitability and limitations of the proposed methods can be studied.B.Method 1The scheme of Fig.3would not directly work for a PR controller as stopping of integrator will introduce large DC offset into controller output.Fig.9represents an anti-windup scheme where the resonant part is withdrawn from the control loop and the integrator outputs are forced to zero if the output of controller goes out of desired range.This can be considered reasonable in situations where feed-forward terms and proportional terms are responsible for a large portion of the control action and the integral terms are responsible only to zero out long term output error.The results are shown in Fig.10.It can be observed that due to the proportional part the output from the controller will not be zero.Reappointing the controller has to be done according to the application,from a higher level master control system,or a manual reset can be a solution for this method.In general to minimize the burden on controller the grid voltage is added as a feed-forward term.The desired range of controller output can be derived from the steady state drop of voltage across the filter between inverter and grid,in case of a grid connected application.Now ifFig.9.PR controller with anti-windup scheme of‘Method1’.Fig.10.Output of a PR controller implemented in FPGA with anti-windup scheme using‘Method1’.Here K p=0.8,K i=125,and ω=314rad/sec.Here Y max=2.5V and Y min=−2.5V.the error is very large and hence the contribution from proportional part becomes significantly high then entire controller output should be forced to zero.C.Method2The second scheme shown in Fig.11uses the dif-ference of actual output y and saturated output y′to be used as feed back signal through a gain K lim.As the saturated output y′would bea clipped sinusoidal wave so feed back through K lim would contain higher order harmonics.But the resonant part of the controller provides low gain other than the resonant frequency. Therefore it can be expected that the output y would not be distorted.However higher value of K lim ensures that output y is small.So,value of K lim can be selected based upon desired range of y for maximum error input e.If value of K lim is high then the magnitudes of harmonics also get increased which in turn distorts the output y.Again for a lower magnitude error input e the Fig.11.PR controller with anti-windup scheme of‘Method2’. saturated output y′follows the output y for longer time.Fig.12.Output of a PR controller implemented in FPGA with anti-windup scheme using‘Method2’.Here K p=0.8,K i=125, K lim=10andω=314rad/sec.Here Y max=2.5V and Y min=−2.5V.The result shown in Fig.12shows that the saturated output is almost following the unsaturated output.Here the input value taken is0.1pu.So for lower input value the output y can be used asfinal output but for higher input value output y becomes significantly distorted. Also if the saturation level chosen are smaller such as10%of rated voltage then level of distortion would increase.IV.E XPERIMENTAL R ESULTTo verify the operation of PR controller a3phase4 wire test setup of inverter with resistive-inductive load was chosen.The neutral point of the load was connected to the dc bus mid point.A current command of4A r.m.s. was given as the input to the current controller.The overall control strategy contains three individual current controller for each phase.The result in Fig.13shows that the controller is capable of tracking the reference current.During the experiment DC bus was held constantat 600V and the load was series connection of of 35Ωresistance and 84mH inductance in each phase.Fig.13.Phase current tracking using the proposed PR controller.Here K p =1.0,K i =1000,K lim =10and ω=314rad/sec.Here Y max =5.0V and Y min =−5.0V .V.C ONCLUSIONThe present paper tries to address the windup or rollover problem that can occur in a PR controller.Two anti-windup schemes have been proposed for PR controller on the basis of the understanding on anti-windup schemes for PI controller.It is shown that a direct extension of the anti-windup mechanism is not feasible.The proposed methods for PR anti-windup implementation indicate that it is possible to obtain an approximate anti-windup performance.The controller logic has been implemented in hardware and results are shown.R EFERENCES[1] C.Bohn and D.P.Atherton,”An analysis package comparingpid anti-windup strategies,”IEEE systems magazine,15(2):34-40,April 1995.[2]H.B.Shin,”New anti-windup PI controller for variable speedmotor drives,”IEEE Transactions on Industrial Electronics,V ol.45,No.3,Page(s)445-450,June 1998.[3] A.Visioli,”Modified anti-windup schemes for pid controllers,”IEE Control Theory and Application,150(1):49-54,January 2003.[4]M.P.Kazmierkowski and L.Malesani,”Current control tech-niques for three-phase voltage-source PWM converters:a sur-vey,”in Industrial Electronics,IEEE Transactions on,V olume 45,Issue 5,Oct 1998Page(s)691-703.[5] A.Scottedward Hodel and C.E.Hall,”Variable-structure pidcontrol to prevent integrator windup,”IEEE Transactions on Industrial Electronics,48(2):442-451,April 2001.[6] D.N.Zmood and D.G.Holmes,”Stationary frame currentregulation of PWM inverters with zero steady-state error,”Power Electronics,IEEE Transactions on V olume 18,Issue 3,May 2003Page(s):814-822.[7]Y .Xiaoming and W.Merk,”The non-ideal generalised amplitudeintegrator (NGAI):interpretation,implementation and applica-tions,”Power Electronics Specialist Conference,2001.PESC.2001IEEE 32nd Annual V olume 4,17-21June 2001Pages(s):1857-1861.Anirban Ghoshal received B.E.degree in Electrical Engineering from the Bengal En-gineering College,Shibpur and M.Tech degree from the Indian Institute of Technology,Delhi.Currently he is working towards his Phd at the Indian Institute of Science,Bangalore.His areas of interests include power electronics,distributed generation anddrives.Vinod John received B.Tech.degree in Elec-trical Engineering from the Indian Institute of Technology,Madras,M.S.E.E.degree from the University of Minnesota,Minneapolis,and Ph.D.from the University of Wisconsin,Madi-son.He has worked in research and de-velopment positions at GE Global Research,Niskayuna,NY and Northern Power,VT.He is currently working as an Assistant Professorat the Indian Institute of Science,Bangalore.His primary areas of interests are in power electronics and distributed generation,power quality,high power converters and motor drives.。