SDRAM 类高速器件布线规则
SDRAM布线规则及技巧
SDRAM布线规则及技巧SDRAM（Synchronous Dynamic Random Access Memory）是一种同步动态随机存取存储器，广泛应用于计算机系统中。
SDRAM的性能受到布线的影响，因此在设计SDRAM布线时需要遵守一些规则和技巧，以保证其性能和稳定性。
首先，布线规则是确保时序要求满足的基础。
时序要求是指控制信号在SDRAM上正确传输和演变的时间约束。
布线规则的主要目标是减少时序延迟和时序失真，确保信号的准确到达。
一些常见的布线规则包括：1.长度匹配：确保信号线的长度尽量相等，可以通过差分对和匹配长度来实现。
2.延迟匹配：保证信号线的延迟尽量相等，可以通过使用等长路径和准确的电缆长度来实现。
3.信号间隔：确保信号线之间的间隔足够，避免相互干扰。
4.层间切换：尽量减少信号线在不同层之间的切换，减少反射和串扰。
其次，布线技巧可以帮助优化布线效果和避免一些常见的布线问题。
1.差分对布线：差分对布线是一种常见的布线技巧，用于减少信号线之间的串扰和提高抗干扰能力。
差分对布线需要保持差分对的长度和延迟匹配。
2.绕行规则：在布线时需要避免绕行，尽量使用直线布线路径，减少信号线的长度和延迟。
3.层交叉规则：避免信号线在不同层之间频繁切换，减少反射和串扰。
如果需要交叉布线，应尽量避免延迟不匹配。
4.电源和地线：为SDRAM提供稳定的电源和地线是非常重要的。
布线时需要确保电源和地线的可靠连接、低噪声和低电阻。
最后，还有一些其他的技巧可以帮助优化SDRAM布线：1.参考设计：可以参考已有的成熟设计和布局规则，避免重复工作和一些常见的布线问题。
2.仿真与验证：使用电磁仿真工具和布线验证工具进行仿真和验证，确保布线符合设计要求并满足性能要求。
3.信号完整性分析：通过信号完整性分析工具，检测和分析信号线中的时序失真和耦合问题，从而优化布线布局。
总结起来，SDRAM布线规则和技巧可以帮助优化布线效果、提高信号完整性，确保SDRAM的性能和稳定性。
SDRAM 等长布线总结
SDRAM 等长布线总结(注：以下内容来自网络，正确与否请自行判断)等长布线总结等长线是为了减少信号相对延时，常用在高速存储器的地址和数据线上，简单来说：等长线的作用，就是让信号传输的速度一致。
I2C总线无需画等长线，虽然i2C信号与内存一样都是有相对时序要求，但由于信号频率较低，此时由导线长度引起的延时不足以影响正常时序，所以无需等长。
当然如果等长也没什么不好的。
差分线与一样，高速信号要注意等长，比如USB，低速信号无需特别注意线长度，比如485。
差分线在布线时要2根线要尽可能保持等距。
高速信号有效的建立保持窗口比较小，要让数据和控制信号都落在有效窗口内，数据、时钟或数据之间、控制信号之间的走线长度差异就很小。
具体允许的偏差可以通过计算时延来得到。
其实一般来说，时序逻辑信号要满足建立时间和保持时间并有一定的余量。
只要满足这个条件，信号是可以不严格等长的。
然而，实际情况是，对于高速信号来说(例如DDR2、DDR3、FSB)，在设计的时候是无法知道时序是否满足建立时间和保持时间要求(影响因素太多，包括芯片内部走线和容性负载造成的延时差别都要考虑，很难通过计算估算出实际值)，必须在芯片内部设置可控延时器件(通过寄存器控制延时)，然后扫描寄存器的值来尝试各种延时，并通过观察信号(直接看波形，测量建立保持时间)来确定延时的值使其满足建立时间和保持时间要求。
不过同一类信号一般只对其中一根或几根信号线来做这种观察，为了使所有信号都满足时序要求，只好规定同一类信号走线全部严格等长。
上面说的是高速并行信号。
对于高速的串行信号，如果是带时钟的，时钟和串行数据也必须满足建立保持时间要求，所以也要控制好长度。
有些高速串行信号虽然带时钟，但这个时钟不是用来锁存数据而是一个频率较低的参考时钟，那么数据和时钟以及多个通道之间的数据的skew就可以宽松很多，不用严格等长，因为接收芯片是能够正确找出每个通道的起始位并且把参考时钟经过PLL倍频和相移来锁存数据的。
sdram pcb 设计规则
sdram pcb 设计规则SDRAM PCB设计规则SDRAM（Synchronous Dynamic Random Access Memory）是一种常见的内存芯片，它在计算机系统和其他电子设备中广泛使用。
为了确保SDRAM能够稳定、高效地工作，PCB（Printed Circuit Board）的设计需要遵循一些特定的规则。
1. 电源和地线规则：SDRAM需要稳定的供电和有效的接地。
为了实现这一点，在设计过程中应遵循以下规则：- 为SDRAM芯片提供一组独立的电源和地线，以减少电源噪声。
- 在布局过程中，将电源和地线尽可能靠近SDRAM芯片，并使用足够宽度的铜层来降低电阻和电感。
2. 布局规则：好的布局是确保SDRAM性能的重要因素。
以下是一些建议的布局规则：- 将SDRAM芯片放置在离CPU和其他重要器件尽可能近的位置，以缩短信号路径。
- 尽量避免将SDRAM芯片放置在热源附近，以防止温度升高而导致性能下降。
- 在布局过程中，遵循良好的信号完整性原则，如避免过长的导线或者过多的弯曲。
3. 信号完整性规则：为了保证信号在PCB上的传输完整性，应遵循以下规则：- 确保时钟和数据线的匹配长度，以防止时序偏差。
- 使用适当的信号层和层间间距来隔离敏感的时序信号，以减少噪声干扰。
- 使用合适的阻抗匹配来提高信号传输的质量，防止反射和信号衰减。
4. 热管理规则：SDRAM在高频运行时会产生热量，因此在设计过程中需要考虑热管理问题：- 在PCB中加入散热孔或金属散热片，以增加散热表面积并提高散热效果。
- 确保周围环境的通风良好，避免过热影响SDRAM的性能。
综上所述，设计符合SDRAM的PCB需要遵循电源和地线规则，布局规则，信号完整性规则以及热管理规则。
这些规则旨在最大程度地提高SDRAM的性能、稳定性和可靠性。
在设计过程中，请确保严格遵守相关规范和标准，以确保SDRAM的最佳工作状态。
SDRAM走线仿真
SDRAM 信号完整性分析SDRAM在车载视觉核心板布线中是最关键的一环，需要慎重考虑其具体的布线策略。
此部分工作频率高，数据吞吐量大，且容易受到外界环境的干扰，也容易影响到其它电气设备的工作，如果数据传输因此出现错误则可能导致严重后果。
一、SDRAM走线规则设定考MICRON提供的关于SDRAM走线建议并根据实际情况对规则设定如下：1、时钟线、控制线、地址线、数据线进行分组，其中数据线和地址线由于线数较多，又分为几个小组（四位一组）；2、设定时钟线、控制线、地址线和数据线的线宽，参考MICRON提供的技术资料，都设定为7mil；3、设定各信号组组内和组间间距，组内保持至少14mil的间距（含与过孔间的间距），组间保持20mil以上的间距；4、相邻两层之间走线尽量保持垂直，减小重合面积，从而减小两线之间的串扰；5、关于去耦电容及SDRAM供电电源引脚走线均采用20mil线宽；6、去耦电容及供电电源脚过孔大小统一设定为内径10mil，外径20mil；二、SDRAM走线仿真规则设定1、驱动时钟的设定驱动时钟根据信号分组中不同信号的工作频率设定不同的驱动时钟周期以及占空比，其中占空比统一设置为50%，地址线组和数据线组的驱动时钟周期统一设定为100MHz，时钟周期也设定为100MHz；2、关于上升沿过冲的设定参考SDRAM 数据手册上关于供电电压和I/O口输入电压的最大电压值（对地4.6V）可以知道允许的最大过冲电压为1.3V，超过该电压很可能导致器件的损坏；3、关于上升沿下冲的设定该值需要参考最小输入高电平电压（2.000V）值，设定为1.3V；4、关于下降沿过冲的设定由于I/O输入最大电压是参考VSS而言，因此也设定为1.3V；5、关于下降沿下冲的设定此处参考SDRAM的IBIS模型中关于阈值电平的电压值1.5V，设定下降沿下冲的最大值为1.5V；6、关于阻抗设定由于DSP和SDRAM的端口特性阻抗皆为50ohm，因此原则上讲应当设定PCB走线的特性阻抗为50ohm，但考虑到电路板的实际情况，设定为40~60ohm；三、布线及仿真中遇见的问题1、关于阻抗匹配的问题由于PCB走线的特征阻抗与其本身的宽度、厚度以及周围的介质有关系，通常带状线和为带线的阻抗也不尽相同，按照目前的PCB层叠结构以及所选用的铜箔的厚度以及填充介质等因素，参考DSP及SDRAM输入输出端口的特征阻抗50ohm，通过软件Polar Si6000计算要使PCB走线的特征阻抗也为50ohm，这需要使线宽约为15.6mil，显然不太合适；考虑降低芯板的厚度来减小走线的宽度，要达到50ohm的特征阻抗，参考欣豐卓群科技（北京）有限公司提供的关于芯板及填充介质的常用厚度，在芯板厚度为4.7mil的情况下，为带线的特性阻抗在线宽为7mil 时能够满足要求；但考虑到电路板是安装在汽车上，汽车会有剧烈的振动，因此如果PCB 板太薄，机械强度很可能不够同时也可能因为产生较大的形变从而使电路板上元器件按损坏导致系统的崩溃，因此拟采用外加匹配电阻的方式来实现阻抗匹配，从而降低反射；2、关于匹配电阻放置位置的问题由于DSP 与SDRAM 之间存在做数据的双向传输，因此严格来说各自都是源端同时也是终端，串联电阻放置在哪段都可以。
DDRSDRAM布线规则
DDRSDRAM布线规则DDRSDRAM布线规则是指在电路板上设计和布置DDRSDRAM的电路和连线时需要遵循的一些规则和原则。
DDRSDRAM是一种双倍速率同步动态随机存储器，用于高速数据存储和访问，因此布线规则尤为重要，可以确保信号的完整性和稳定性，提高系统的性能和可靠性。
以下是DDRSDRAM布线规则的一些重要方面：1.线长匹配：DDRSDRAM的布线中，所有的时钟、地址、数据和控制信号必须尽量保持相等的线长。
由于DDRSDRAM使用双倍速率，信号频率较高，线长差异可能导致信号到达时间不一致，影响系统的稳定性。
通过保持线长相等，可以降低信号的传输延迟，减少时钟失真和时序错误。
2.地与电源平面：DDRSDRAM的布线中，要为信号线和电源线提供良好的地和电源环境。
通过使用地和电源平面，可以降低信号线上的互损耗和串扰，提高信号的信噪比和阻抗匹配。
电源平面还可以提供稳定的电源供应，减少功率噪声和波动对信号传输的影响。
3.信号隔离：DDRSDRAM的布线中，需要将不同类型的信号线进行隔离，避免互相干扰。
例如，时钟信号和数据信号应尽量分开布线，以减少互相之间的串扰。
同时，还应将高速信号线和低速信号线进行分离，避免高速信号对低速信号的影响。
4.差分信号：DDRSDRAM的部分信号采用差分传输方式，例如，地址和数据线。
在布线时，要确保差分线对称和匹配。
差分线对称性可以减少共模噪声的影响，而差分线匹配可以提高差分信号的传输效率和抗干扰能力。
5.终端电阻：DDRSDRAM的布线中，需要正确设置终端电阻来匹配信号线的特性阻抗。
终端电阻的作用是反射信号的能量，减少信号反射和回波干扰。
正确设置终端电阻可以提高信号的传输质量，减少时序错误和噪声。
6.时序调整：DDRSDRAM的布线中，需根据具体的DDRSDRAM芯片和系统要求进行时序调整。
时序调整包括延迟设置、预充电设置和时钟节拍调整等。
通过合理设置时序参数，可以确保DDRSDRAM正常工作，提高数据传输的稳定性和速度。
DDR2 DDR3 SDRAM的PCB布线规则指导
Signal Integrity and PCB layout considerations for DDR2-800 Mb/s and DDR3 MemoriesFidus Systems Inc.900, Morrison Drive, Ottawa, Ontario, K2H 8K7, CanadaChris Brennan, Cristian Tudor, Eric Schroeter, Heike Wunschmann, and Syed BokhariSession # 8.13AbstractThe paper addresses the challenge of meeting Signal Integrity (SI) and Power Integrity (PI) requirements of Printed Circuit Boards (PCBs) containing Double Data Rate 2 (DDR2) memories. The emphasis is on low layer count PCBs, typically 4-6 layers using conventional technology. Some design guidelines have been provided.1. IntroductionDDR2 usage is common today with a push towards higher speeds such as 800 Mbps [1] and more recently, 1066 Mbps. DDR3 [2] targets a data rate of 1600 Mbps. From a PCB implementation standpoint, a primary requirement is delay matching which is dictated by the timing requirement. This brings into it a number of related factors that affect waveform integrity and delay. These factors are interdependent, but where a distinction can be made, they can be termed PCB layer stackup and impedance, interconnect topologies, delay matching, cross talk, PI and timing. Cadence ALLEGRO™SI-230 and Ansoft’s HFSS™ are used in all computations.Table 1: Comparison of DDR2 and DDR3 requirementsSignals common to both technologies and a general comparison of DDR2 and DDR3 is shown in Table 1. It must be noted that “matching” includes cases where the clock net may be made longer (termed DELTA in ALLEGRO SigXP). We have assumed a configuration comprising a Controller and two SDRAMs in most illustrations that follow.2. PCB Layer stackup and impedanceIn a layer constrained implementation, a 4 layer PCB (Figure 1) is a minimum with all routing on TOP and BOTTOM layers. One of the internal layers will be a solid ground plane (GND). The other internal plane layer is dedicated to VDD. Vtt and Vref can be derived from VDD. Use of a 6-layer PCB makes the implementation of certain topologies easier. PI is also enhanced due to the reduced spacing between power and GND planes. The interconnect characteristic impedance for DDR2 implementation can be a constant. A single-ended trace characteristic impedance of 50 Ohms can be used for all single-ended signals. A differential impedance of 100 Ohms can be used for all differential signals, namely CLOCK and DQS. Further, the termination resistor pulled up to VTT can be kept at 50 Ohms and ODT settings can be kept at 50 Ohms.In the case of DDR3 however, single ended trace impedances of 40 and 60 Ohms used selectively on loaded sections of ADDR/CMD/CNTRL nets have been found to be advantageous. Further, the value of the termination resistor pulled up to Vtt needs to be optimized in combination with the trace impedance through SI simulations. Typically, it is in the range 30 – 70 Ohms. The differential trace impedance can remain at 100 Ohms.Figure 1 : Four and Six layer PCB stackup3. Interconnect TopologiesIn both cases of DDR2 and DDR3, DQ, DM and DQS signals are point-to-point and do not need any topological consideration. An exception is in the case of multi-rank Dual In Line Memory Modules (DIMMs). Waveform integrity is also easily addressed by a proper choice of drive strengths and On Die Termination (ODT). The ADDR/CMD/CNTRL signals, and sometimes the clock signal will involve a multipoint connection where a suitable topology is needed. Possible choices are indicated in Figure 2 for cases involving two SDRAMs. The Fly-By Topology is a special case of a daisy chain with a very short or no stub.For DDR3, any of these topologies will work, provided that the trace lengths are minimized. The Fly-by topology shows the best waveform integrity in terms of an increased noise margin. This can be difficult to implement on a4-layer PCB and the need for a 6-layer PCB arises. The daisy chain topology is easier to implement on a 4 layer PCB. The tree topology on the other hand requires the length of the branch AB to be very close to that of AC (Figure 2). Enforcing this requirement results in the need to increase the length of the branches which affects waveform integrity. Therefore, for DDR3 implementation, the daisy chain topology with minimized stubs proves to be best suited for 4-layer PCBs.For DDR2-800 Mbps any of these topologies are applicable with the distinction between each other being less dramatic. Again, the daisy chain proves to be superior in terms of both implementation as well as SI.Where more than two SDRAMs are present, often, the topology can be dictated by constraints on device placement. Figure 3 shows some examples where a topology could be chosen to suit a particular component placement. Of these, only A and D are best suited for 4-layer PCB implementation. Again, for DDR2-800 Mbps operations all topologies yield adequate waveform integrity. For a DDR3 implementation, in particular at 1600 Mbps, only D appears to be feasible.Vtt RtRtRtTree topology Fly-By topologyFigure 2: ADDR/CMD/CNTRL topologies with 2 SDRAMS(A)Figure 3: ADDR/CMD/CNTRL topologies with four SDRAMS4. Delay matchingImplementing matched delay is usually carried out by bending a trace in a trombone shape. Routing blockage may require layer jumping. Unfortunately, while physical interconnect lengths can be made identical in layout, electrically, the two configurations shown in Figure 4 will not be the same.The case of trombone delay has been well understood, and the case of a via is obvious. The delay of a trombone trace is smaller than the delay of a straight trace of the same center-line length. In the case of a via, the delay is more than that of a straight microstrip trace of length equal to the via length. The problem can be resolved in two different ways. In the first approach, these values can be pre-computed precisely and taken into account while delay matching. This would become a tedious exercise which could perhaps be eased with userRtRtRt(B)(C)(D)Rtdefined constraints in ALLEGRO 16.0. In the second approach, one would use means to reduce the disparity to an acceptable level.Trombone traceStraight traceL 3L 2 L 4 ≠L 1L 5Figure 4: Illustration of Trombone traces and ViasFigure 5: Circuit for estimation of trombone effect and resulting waveforms.≠Straight traceVia cross sectional viewConsider the case of a trombone trace. It is known that the disparity can be reduced by increasing the length of L3 (Figure 4). Details can be found in reference [3]. A simulation topology can be set up in SigXP to represent parallel arms of a trombone trace as coupled lines. A sweep simulation is carried out with L3 (S in Figure 5) as a variable and the largest reasonable value that reduces the delay difference with respect to a reference trace is selected. For microstrip traces, L3 > 7 times the distance of the trace to ground is needed.Delay values are affected in a trombone trace due to coupling between parallel trace segments. Another way to reduce coupling without increasing the spacing is to use a saw tooth profile. The saw tooth profile shows better performance as compared to a trombone although it eventually ends up requiring more space. In either case, it is possible to estimate the effect on delay precisely by using a modified equation for the computation of the effective trace length [3]. This would need to be implemented as a user defined constraint in ALLEGRO.Consider the case of a through hole via on the 6 layer stackup of Figure 2. Ground vias placed close to the signal vias play an important role in the delay. For the illustration, the microstrip traces on TOP and BOTTOM layers are 150 mils long, and 4 mils wide. The via barrel diameter = 8 mils, pad diameter is 18 mils and the anti-pad diameter is 26 mils.Three different cases are considered. In the first case, the interconnect with via does not have any ground vias in its immediate neighborhood. Return paths are provided at the edges of the PCB 250 mils away from the signal via. In the second case, a reference straight microstrip trace of length = 362 mils is considered. The third case is the same as case 1 with four ground vias in the neighborhood of the signal via. Computed s-parameters with 60 Ohm normalization are shown in Figure 6. It can be seen that the use of 4 ground vias surrounding the signal via makes its behavior more like a uniform impedance transmission line and improves the s21 characteristic. In the absence of a return path in the immediate neighborhood, the via impedance increases. For the present purpose, it is important to know the resulting impact on the delay.A test circuit is set up similar to Figure 5. The driver is a linear source of 60 Ohms output impedance and outputs a trapezoidal signal of rise time = fall time = 100 ps and amplitude = 1V. It is connected to each of the 3 interconnects shown in Figure 6 and the far end is terminated in a 60 Ohm load. The excitation is a periodic signal with a frequency of 800 MHz. The time difference between the driver waveform at V = 0.5 V and the waveform at the receiver gives the switched delay.Results are illustrated in Figure 7 where only the rising edge is shown. It can be seen that the delay with four neighboring ground vias differs from that of the straight trace by 3 ps. On the other hand, the difference is 8 ps for the interconnect with no ground vias in the immediate neighborhood.It is therefore clear that increasing the ground via density near signal vias will help. However, in the case of 4 layer PCBs, this will not be possible as the signal traces adjacent to the Power plane will be referenced to a Power plane. Consequently, the signal return path would depend on decoupling. Therefore, it is very important that the decoupling requirement on 4 layer PCBs addresses return paths in addition to meeting power integrity requirements.The clock net is differential in both DDR2 and DDR3. In DDR2, DQS can be either single ended or differential although it is usually implemented as differential at higher data rates. The switched delay of a differential trace is less than that of a single ended trace of identical length. Where timing computations indicate the need, the clock and DQS traces may need to be made longer than the corresponding ADDR/CMD/CNTRL nets and DATA nets.and DQ nets.Since DQ and DM nets run at the maximum speed, it is desirable that all of these nets in any byte lane be routed identically, preferably without vias. Differential nets are less sensitive to discontinuities and where layer jumping is needed, the DQS and CLOCK nets should be considered first.Figure 6: s-parameters of interconnects with vias (60 Ohm normalization)Figure 7: Driver and Receiver waveforms for the 3 cases of Figure 6. (Plot colors correspond)5. CrosstalkCross talk contributes to delay uncertainty being significant for microstrip traces. This is generally reduced by increasing the spacing between adjacent traces for long parallel runs. This has the drawback of increasing the total trace length and therefore a reasonable value must be chosen. Typically the spacing should be greater than twice the trace distance to ground. Again, ground vias play an important role. Near and far end coupling levels are illustrated in Figure 8. Use of multiple ground vias reduces coupling levels by 7 dB. To derive the interconnect budget, a simulation of a victim trace with two aggressors on both sides is adequate. Using a periodic excitation on all nets will yield the cross talk induced jitter. Using a pseudo random excitation on all nets will show the effect of both cross talk as well as data dependencies. Time domain results are not shown here, but it is easily done by setting up a 5 coupled line circuit in SigXP with the spacing between traces set up for sweeping. Reasonable spacing values that keep the jitter in the waveform due to both cross talk as well as pattern dependence at an acceptable level are chosen.Figure 8: s-parameters of coupled traces (60 Ohm normalization)6. Power IntegrityPower Integrity here refers to meeting the Power supply tolerance requirement under a maximum switching condition. Failure to address this requirement properly leads to a number of problems, such as increased clock jitter, increased data dependent jitter, and increased cross talk all of which eventually reduce timing margins.The theory for decoupling has been very well understood and usually starts with the definition of a “target impedance” as [4]CurrentTransient tolerance Voltage Z et t =arg (1)An important requirement here is knowledge of the transient current under worst case switching condition. A second important requirement is the frequency range. This is the range of frequencies over which the decoupling network must ensure that its impedance value is equal to or below the required target impedance. On a printed circuit board, capacitance created by the Power-Ground sandwich and the decoupling capacitors needs to handle a minimum frequency of ~100 kHz up to a maximum frequency of ~100-200 MHz. Frequencies below 100 kHz are easily addressed by the bulk capacitance of the voltage regulator module. Frequencies above 200 MHz should be addressed by the on-die and in some cases on-package decoupling capacitance. Due to the finite inductance of the package, there is no need to provide decoupling on the PCB to handle frequencies greater than 200 MHz. The actual computation of power integrity can be very complex involving IC package details, simultaneously switched signals and the PCB power distribution network. For PCB design, the use of the target impedance approach to decoupling design is simpler and provides a practical solution with very little computational effort.The three power rails of concern are the VDD, VTT and Vref. The tolerance requirements on the VDD rail is ~ 5% and the transient current is determined as the difference between Idd7 and Idd2 as specified by JEDEC [1,4]. This is accomplished by using plane layers for power distribution and a modest number of decoupling capacitors. It is preferable to use decoupling capacitors of 10 different values distributed in the range of 10 nF to 10 uF. Further, the capacitor pad mounting structure should be designed for reduced mounted inductance.The Vref rail has a tighter tolerance, but it draws very little current. Its target impedance is easily met using narrow traces and one or two decoupling capacitors. It is important however that the capacitors be located very close to the device pins.The VTT rail proves to be challenging because it not only has a tighter tolerance, but it also draws a transient current close to that of the VDD rail. The transient current is easily calculated as described in reference [5]. Again, the target impedance requirement can be met using an increased number of decoupling capacitors.On a 4 layer PCB, the planes are too far apart and consequently the advantage of inter-plane capacitance is lost. The number of decoupling capacitors needs to be increased and higher frequency capacitors with values less than 10 nF may be needed. These computations are easily done using ALLEGRO SI Power Integrity option.7. TimingTiming computation is carried out as described in reference [6]. A table needs to be setup for the following eight cases: 1. 2. 3. 4. 5. 6. 7. 8. Write Setup analysis DQ vs. DQS Write Hold analysis DQ vs. DQS Read Setup analysis DQ vs. DQS Read Hold analysis DQ vs. DQS Write Setup analysis DQS vs. CLK Write Hold analysis DQS vs. CLK Write Setup analysis ADDR/CMD/CNTRL vs. CLK Write Hold analysis ADDR/CMD/CNTRL vs. CLKAn example is shown for the case of Write setup analysis in Table 2. Actual numbers have been omitted as they are not precisely known yet for DDR3. These numbers are obtained from data sheets of Controller and memory manufacturers. The numbers in the interconnect section are determined by SI simulations. All the eight cases need to be analyzed for DDR2. For DDR3, 5 and 6 are not needed due to its write leveling feature. In the PCB implementation, length match tolerances must ensure that the total margin is positive. ElementControllerSkew Componenta.)DQ vs. DQS skew at transmitter output b.) Data / Strobe PLL jitter a+b Setup requirement (tDSb @ Vih/Vil level) DQ slew rate DQS slew rateSetupUnitsps ps ps psCommentsFrom controller design data Used if not included in transmitter skewTotal Controller SDRAM (or DIMM)V/ns V/ns psTotal SDRAM setup requirement InterconnecttDSb + slew rate adjustmentFrom SDRAM datasheet; this number is to be adjusted based on DQ and DQS slew rates Measured as per JEDEC specification from SI simulation results Measured as per JEDEC specification from SI simulation results Includes slew rate adjustmenta.) Data Xtalk b.) DQS Xtalk c.) Length matching tolerance d.) Characteristic impedance mismatch Total Interconnect Min. Total Setup Budget Setup margin Interconnect skew (a + b + c + d) 0.24*tckps ps ps ps2 aggressors (one each side of the victim); victim – repetitive; aggressor- PRBS 2 aggressors (one each side of the victim); victim – repetitive; aggressor- PRBS Extracted from SI simulation results longest data net, worst case PVT corner can be omitted if routing of DQ and corresponding DQS signals are done on same layerps ps From SDRAM datasheet (includes clock duty cycle variation) Must be positiveMin. Total Setup Budget – (Total Controller + Total SDRAM + Total Interconnect )psTable 2: Illustration of DDR3 Write Setup timing analysis summary for DQ vs. DQS8. PCB LayoutImplementation on a PCB involves a number of tradeoffs to meet SI requirements. Often, the question is how far does one need to go? PCB layout tasks are facilitated using the following approach: 1. Set up topology and constraints in ALLEGRO Constraint Manager. 2. Design Controller BGA breakout. A controller pin arrangement with ADDR/CMD/CNTRL pins in the middle and DQ/DQS/DM byte lanes on either side is best suited. Within these groups, individual pins may need to be swapped to ensure routing with minimum cross-over. 3. Attempt routing with reduced stub length and a minimum trace spacing as obtained from cross talk simulation. Often, most stubs can be eliminated but it will not be possible for all the pins. One may try two traces between BGA pads of the memory devices. This would require narrow PCB traces which can increase manufacturing cost. Yet, it will not be possible for all signals unless micro via and via-in-pad technology is used. Complete routing with coarse length matching tolerances. 4. Place Vref decoupling capacitors close to the Vref pins. Vtt decoupling can be placed at the far end of the last SDRAM and will not come in the way of routing. VDD decoupling can be placed close to devices where possible without blocking routing channels. The smaller valued capacitors should be placed closer to the devices. With a proper decoupling design, it will not be necessary to cram all capacitors close to the devices. All decoupling capacitors should use a fan out for the footprint designed for reduced inductance. This is typically two short wide traces perpendicular to the capacitor length. This can be automated by using a user defined capacitor footprint that can be attached to all the decoupling capacitors in the schematic. 5. Implement fine length matching and insert multiple ground vias where signal traces jump layers. It is better to use the delay matching option in ALLEGRO and one must include z-axis delay. Typically, P and N nets of differential pairs should be matched with a tolerance of +/- 2ps and the tolerance for all other matched nets can be +/- 10 ps or more based on the timing margin computation.9. DIMMConsiderations described above apply to the case of PCBs containing one or more DIMMs. The only exception is that the decoupling requirement for the memories can be relaxed as it is already accounted for on the DIMM PCB. SI analysis of registered DIMMs is also much simpler where the DIMM is treated as a single load. While the routing topology for ADDR/CMD/CNTRL nets is usually a daisy chain with reduced stubs, tree topologies can also be used for registered DIMMs. Analysis of un-buffered DIMMs can become tedious as the timing requirement at all the SDRAMs must be analyzed. DIMM routing on 4-layer PCBs is relatively simpler compared to the case of SDRAMs.10. ExamplesThe detail described above has been used in the implementation of a DDR2 PCB, a DDR3 PCB and a DDR3 – DIMM PCB. The controller is from MOSAID [7] which is designed to provide both DDR2 as well as DDR3 functionality. For the SI simulations, IBIS models have been used. Models for the memories are from MICRON Technology, Inc [8]. The IBIS models for the DDR3 SDRAMs were available at 1333 Mbps speed. These were used at 1600 Mbps. For the unbuffered DDR3 DIMM (MT_DDR3_0542cc) EBD models from Micron Technology were used. All waveforms are for the typical case and are computed at the SDRAM die. The 6 layer PCB stackup of Figure 2 is used with routing on TOP and BOTTOM layers only. The memory consists of 2 SDRAMsrouted as a daisy chain. In the case of the DIMM, a single unbufferred DIMM is used. TOP/BOTTOM layer routing and Signal Integrity waveforms are shown in Figures. 9-11.Snapshots ofFigure 9: Illustration of TOP and BOTTOM layers of a DDR3 PCB with computed waveforms at the farthest SDRAM. Waveform on left is an ADDRESS net compared to that of the CLOCK net. Waveform on the right is a DATA net compared to that of a DQS net. Clock frequency = 800 MHz and data rate is 1600 Mbps.Figure 10: Illustration of TOP and BOTTOM layers of a DDR2 PCB with computed waveforms at the farthest SDRAM. Waveform on left is an ADDRESS net compared to that of the CLOCK net. Waveform on the right is a DATA net compared to that of a DQS net. Clock frequency = 400 MHz and data rate is 800 Mbps.Figure 11: Illustration of TOP and BOTTOM layers of a DDR3 – DIMM PCB with computed waveforms at the 8th (last) SDRAM on DIMM. Waveform on left is an ADDRESS net compared to that of the CLOCK net. Waveform on the right is a DATA net compared to that of a DQS net.Lastly, Figure 12 shows a comparison of computed and measured DATA eye patterns of an 800 Mbps DDR2. In all cases waveform integrity can be seen to be excellent.Figure 12: Computed (Red) and Measured (blue) waveforms of a data net of an 800 Mbps DDR2 PCB.11. ConclusionIn this paper, all aspects related to SI, and PI of DDR2 and DDR3 implementation have been described. Use of Constraint Manager in ALLEGROTM makes implementation easy. While a four layer PCB implementation of 800 Mbps DDR2 and DDR3 appears to be feasible, DDR3-1600 Mbps will prove to be challenging. It will become clearer as the memory devices become available and one has a good handle on timing numbers.References[1] DDR2 SDRAM Specification, JEDEC JESD79-2B, January 2005. [2] DDR3 SDRAM Standard, JEDEC JESD79-3, June 2007. [3] Syed Bokhari, “Delay matching on Printed Circuit Boards”, Proceedings of the CDNLIVE 2006, San Jose. [4] Larry D Smith, and Jeffrey Lee, “Power Distribution System for JEDEC DDR2 memory DIMM, Proc. IEEE EPEP conference, Princeton, N.J., pp. 121-124, October 2003. [5] Hardware and layout design considerations for DDR2 SDRAM Memory Interfaces, Freescale semiconductor Application Note, Doc. No. AN2910, Rev. 2, 03/2007. [6] DDR2 design guide for 2 DIMM systems, Technical Note, Micron Technology Inc. TN-47-01, 2003. [7] /corporate/products-services/ip/SDRAM_Controller_whitepaper_Oct_2006.pdf [8] /products/dram/ddr2/partlist.aspx?speed=DDR2-800 [9] /products/dram/ddr3/partlist.aspx?speed=DDR3-1066。
sdram pcb 设计规则
sdram pcb 设计规则SDRAM (Synchronous Dynamic Random Access Memory)是一种同步动态随机存取存储器，被广泛应用于计算机和其他电子设备中。
SDRAM PCB (Printed Circuit Board)设计规则是在设计SDRAM模块时需要遵循的一些准则和规范，以确保模块的性能、稳定性和可靠性。
下面是一些与SDRAM PCB设计相关的参考内容：1. PCB布局：- 确保SDRAM芯片和相关元件之间的连接尽可能短，以减少信号传输的延迟。
- 高速信号线应避免过长的走线，并尽量采用直线走线方式，以减少信号的反射和干扰。
- 将SDRAM芯片和电源引脚放置在接近功耗滤波电容的位置，以最大程度地降低功耗线的阻抗。
- 合理规划地面和电源平面，确保它们之间有足够的距离，以减少地平面与电源平面之间的串扰。
2. 信号完整性：- 为时钟信号、地址信号和控制信号提供低阻抗、低噪声的电源电压。
这可以通过增加电源滤波电容和合理布局电源和地线来实现。
- 使用阻抗匹配技术，保持信号走线的阻抗与适配SDRAM的驱动器和终端之间的要求一致。
- 通过添加补偿差分走线、增加差分走线间距、使用扇出缓冲器等措施，减少信号串扰和互相干扰。
3. 电源和地线：- 提供足够的地平面和电源平面，以减少信号回流路径的长度和电磁干扰。
- 采用较大的电源与地引脚走线，以增加电源回流的路径，减小引脚区距离，提高电源稳定性。
- 使用分区式供电和分离式地线布局，以降低供电噪声和信号引起的传导和射频辐射干扰。
4. DDR引脚布局和输形：- DDR (Double Data Rate)是SDRAM的一种改进版本，它有更高的数据传输速率和更复杂的信号分布。
在布局和输形过程中，应遵循DDR的特殊要求，如匹配长度差异、避免信号回流突变等。
5. 噪声控制：- 在PCB设计中使用分离式地线和电源布局可减少地线回流并降低供电噪声。
sdram信号线等长处理
sdram信号线等长处理SDRAM（同步动态随机存取存储器）是一种常见的内存类型，被广泛应用于计算机和其他电子设备中。
在设计电路板时，处理SDRAM信号线等长是非常重要的。
如果SDRAM信号线长度不一致，可能会导致信号传输延迟，影响系统性能。
那么，如何处理SDRAM信号线等长呢？首先，我们需要了解SDRAM信号线的作用。
SDRAM信号线用于在主板和SDRAM芯片之间传输数据和控制信号。
由于信号在传输过程中会遇到阻抗不匹配和传输延迟等问题，因此需要将SDRAM信号线等长处理，以保证信号的同步和稳定性。
在处理SDRAM信号线等长时，首先要进行信号路径规划。
即确定信号线的走向和长度，使各个信号线的长度尽量保持一致。
这样可以降低信号传输延迟，提高系统的稳定性和性能。
在进行信号路径规划时，可以使用专业的设计软件，如Cadence Allegro等，来帮助实现信号线等长处理。
另外，还可以采用控制线的方式来处理SDRAM信号线等长。
即引入控制线，通过调整控制线的长度来实现信号线的等长处理。
这种方法可以有效地控制信号线的长度差异，提高系统的稳定性和可靠性。
除了信号路径规划和控制线调整外，还可以采用差分信号线和匹配线的方式来处理SDRAM信号线等长。
差分信号线可以有效地抑制信号干扰和传输失真，提高系统的抗干扰能力。
而匹配线则可以确保信号线的长度一致，提高系统的同步性和稳定性。
总的来说，处理SDRAM信号线等长是设计电路板时需要重点考虑的问题。
通过合理的信号路径规划、控制线调整、差分信号线和匹配线等方法，可以有效地实现SDRAM信号线的等长处理，提高系统的性能和可靠性。
希望以上内容能对大家有所帮助，谢谢！。
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Learn to walk first before you want to run…SDRAM 类高速器件布线规则
一个优秀的Layout，一块好的板子，并不是随便布线连同就可以实现电路要求的，凡事都得谨慎，此处别处摘要，讲述SDRAM类高速器件布线规则：
如果你没有信号完整性的知识和对传输线的认识，恐怕你很难看懂，如果你看不懂，那么请按这样一个通用的基本法则做：
（1）DDR和主控芯片尽量靠近
（2）高速约束中设置所有信号、时钟线等长(最多允许50mils的冗余)，所有信号、时钟线长度不超过1000mils
（3）尽量0过孔，元件层下面一定要有一个接地良好的地层，所有走线不能跨过地的分割槽，即从元件层透视地层看不到与信号线交叉的地层分割线。
这样的话200M的DDR基本上是没有太大问题。
其它的一些3W 20H法则就能做到尽量做到吧
3W原则:
这里3W是线与线之间的距离保持3倍线宽。
你说3H也可以。
但是这里H指的是线宽度。
不是介质厚度。
是为了减少线间串扰，应保证线间距足够大，如果线中心距不少于3倍线宽时，
则可保持70%的线间电场不互相干扰，称为3W规则。
如要达到98%的电场不互相干扰，可使用10W规则。
针对EMI（电磁干扰：eg传导、辐射、谐波）
20H原则：
是指电源层相对地层内缩20H的距离，当然也是为抑制边缘辐射效应。
在板的边缘会向外辐射电磁干扰。
将电源层内缩，使得电场只在接地层的范围内传导。
有效的提高了EMC。
若内缩20H则可以将70%的电场限制在接地边沿内；内缩100H则可以将98%的电场限制在内。
针对EMC（电磁兼容）
五---五规则：
印制板层数选择规则，即时钟频率到5MHz或脉冲上升时间小于5ns，则PCB板须采用多层板，这是一般的规则，有的时候出于成本等因素的考虑，采用双层板结构时，这种情况下，最好将印制板的一面做为一个完整的地平面层。
对于“五五规则”的时钟频率到5MHz或脉冲上升时间小于5ns，此处我严重不理解。
时钟信号：以地平面为参考，给整个时钟回路的走线提供一个完整的地平面，
给回路电流提供一个低阻抗的路径。
由于是差分时钟信号，在走线前应预先设计好线宽线距，计算好差分阻抗，再按照这种约束来进行布线。
所有的DDR差分时钟信号都必须在关键平面上走线，尽量避免层到层的转换。
线宽和差分间距需要参考DDR控制器的实施细则，信号线的单线阻抗应控制在50～60 Ω，差分阻抗控制在100～120 Ω。
时钟信号到其他信号应保持在20
mil*以上的距离来防止对其他信号的干扰。
蛇形走线的间距不应小于20 mil。
串联终端电阻RS值在15～33Ω，可选的并联终端电阻RT值在25～68 Ω，具体设定的阻值还是应该依据信号完整性仿真的结果。
数据信号组：以地平面为参考，给信号回路提供完整的地平面。
特征阻抗控制在50～60 Ω。
线宽要求参考实施细则。
与其他非DDR信号间距至少隔离20 mil。
长度匹配按字节通道为单位进行设置，每字节通道内数据信号DQ、数据选通DQS和数据屏蔽信号DM长度差应控制在±25 mil内(非常重要)，不同字节通道的信号长度差应控制在1 000 mil 内。
与相匹配的DM和DQS串联匹配电阻RS值为0～33 Ω，并联匹配终端电阻RT值为25～68Ω。
如果使用电阻排的方式匹配，则数据电阻排内不应有其他DDR信号。
地址和命令信号组：保持完整的地和电源平面。
特征阻抗控制在50～60 Ω。
信号线宽参考具体设计实施细则。
信号组与其他非DDR信号间距至少保持在20 mil
以上。
组内信号应该与DDR时钟线长度匹配，差距至少控制在25 mil内。
串联匹配电阻RS 值为O～33 Ω，并联匹配电阻RT值应该在25～68 Ω。
本组内的信号不要和数据信号组在同一个电阻排内。
控制信号组：控制信号组的信号最少，只有时钟使能和片选两种信号。
仍需要有一个完整的地平面和电源平面作参考。
串联匹配电阻RS值为O～33 Ω，并联匹配终端电阻RT值为25～68 Ω。
为了防止串扰，本组内信号同样也不能和数据信号在同一个电阻排内。