高速车速预警系统方案-CMOS-endversion

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汽车车速检测系统设计概要

3.3 显示模块的选型
目前用于显示的基本有液晶LCD显示板和二极管LED数码管显示。由于此次的车速检测系统构造简单且设计方便，汽车测速显示需要较高的刷新速率才能更加好的实现实时速度显示。所以本次选用的是比较简单的LED数码管来实现速度的显示。
LED数码管显示中分为静态显示和动态显示两种。
静态驱动是指每个数码管的每一个段码都由一个单片机的I/O口进行驱动，或者使用如BCD码二-十进位器进行驱动。静态驱动的优点是编程简单，显示亮度高，缺点是每个数码管需要占用一个锁存器，硬件电路复杂。因为数码管都处于被点亮状态，所以需要的电流很大，当数码管的数量增多时，对电源的要求也就随之增高。
4 系统硬件设计
4.1 AT89C51
AT89C51是一种带4K字节FLASH存储器（FPEROM—Flash Programmable and Erasable Read Only Memory）的低电压、高性能CMOS 8位微处理器，俗称单片机[6]。AT89C2051是一种带2K字节闪存具有可编程可擦除且只读存储器的单片机。51单片机的可擦除和只读存储器能够反复的擦除1000次。该器件是利用ATMEL高密度的非易失的存储器制造技术来制造的，其与工业标准的MCS-51指令集和输出管脚互相兼容。由于把多功能8位CPU以及闪速存储器组合于单个芯片里面，ATMEL的51单片机成为一种高效的微控制器。AT89C51单片机能够提供一种较高灵活性以及价格相对低廉的方案给嵌入式控制系统来使用。
所以本次选用的是LED数码管的动态显示。
3.4 报警电路选择
蜂鸣器是一种一体化结构的电子讯响器，采用直流电压供电，广泛应用于计算机、打印机、复印件、报警器、玩具、汽车电子设备、定时器等的电子产品中，用做发声器件。蜂鸣器主要分为压电式蜂鸣器和电磁式蜂鸣器两种类型。

基于单片机的汽车超速报警器的设计

基于单片机的汽车超速报警器的设计随着社会的发展和科技的进步，汽车已成为人们日常生活的重要交通工具。

然而，不适当的驾驶速度可能导致交通事故和生命财产的损失。

因此，设计一种基于单片机的汽车超速报警器，对保障行车安全具有重要意义。

一、设计背景与意义汽车超速报警器是一种通过监测车辆行驶速度并判断是否超速的装置。

当车辆行驶速度超过设定阈值时，报警器会发出警报，提醒驾驶员减速。

该装置有助于减少因超速驾驶导致的交通事故，提高道路安全。

二、硬件设计1、传感器选择：选用霍尔传感器作为车速传感器，其输出电压与转速成正比，可用于测量汽车行驶速度。

2、单片机选择：采用AT89C51单片机作为核心控制器，该单片机具有低功耗、高性能的特点，满足汽车行驶中的恶劣环境要求。

3、报警装置：采用蜂鸣器和LED灯作为报警装置，当汽车超速时，蜂鸣器发出警报声，LED灯闪烁提示。

4、存储模块：为保存设定的速度阈值和超速记录，需设计一个非易失性存储模块，如EEPROM。

5、电源模块：考虑到汽车电源的特殊性，设计一个稳定的电源模块，以确保报警器的稳定工作。

三、软件设计1、速度采集：通过霍尔传感器采集汽车行驶速度，并将速度信号转换为电信号输入单片机。

2、速度判断：单片机读取速度信号后，与设定的速度阈值进行比较。

若超速，则触发报警装置。

3、报警处理：当报警触发时，单片机控制蜂鸣器发出警报声，LED 灯闪烁提示。

同时，将超速记录保存在存储模块中。

4、速度阈值设定：为适应不同路况和驾驶需求，软件中设计一个速度阈值设定功能，驾驶员可根据实际情况调整阈值。

5、程序优化：为提高程序效率和稳定性，采用模块化设计和中断处理技术，减少CPU的占用时间。

四、系统测试与优化1、速度测试：通过实际行驶测试，验证报警器是否能准确监测汽车速度，并判断是否超速。

2、硬件调试：检查电路板连接是否正确，调整传感器和报警装置的工作状态，确保系统正常运行。

3、软件调试：通过调试和优化程序，提高报警器的响应速度和准确性。

基于单片机的汽车超速警报系统设计-毕业设计任务书

国外研究状况：
国外对超速进行研究开展较早也较为成熟，已经有相关学者建立了各种不同的车速模型。西方发达国家对汽车运行速度的研究主要集中在公路经济分析领域。早在20世纪60年代，西方发达国家就开始对超速行驶进行研究。其中的基础性工作就是对车辆运行速度的研究，并进行了大量的道路试验。
车辆行驶信息采集的设备如美国的Autoscoe系统，是最早获得专利的视频车辆监测系统，可以对道路的车流量，车速等路况信息进行采集。美国采用最高法定限速值和分段限速相结合的方法限速。最高法定限速是美国早期限速的主要方法，适用于美国的所有道路。由于全国道路条件和沿线环境的多样化，同一的法定限速值并不能满足各地道路运行的实际情况，进行分段测速。日本的SEIICHIKAGAYA建立了一个辅助决策系统并应用于Sapporo市，通过对道路环境因素建模，为决策者提供决策信息来提高运输效率。德国的大部分公路不限速；法国采用可变限速，一般晴天和雨天采用不同的限速值；新西兰则根据道路条件、交通量、事故资料或周围环境等因素将道路分为若干交通安全等级，将其作为制定限速值的依据，其限速值的范围为20-100km/h。
Key words:Single chip microcomputer;Automobile overspeed alarm;Voice sensor;The LCD display
摘要
Abstract
1前言
1.1
在当今时代里，汽车时最普遍的交通运输工具，随着汽车工业和告诉公路建设的发展，每年由于各种交通事故造成的人员伤亡数目惊人，给国家造成的经济损失巨大。据统计，每年造成各种交通事故的最重要的原因是车辆的超载和驾驶员的超速行驶，而后者随机性往往更大可是人们总是忽视超速行驶给人们带来的严重后果。每年由于司机超速行驶而造成的交通事故非常多，由此造成的经济损失也很严重。可是很多因超速行驶而造成事故者或许并不是他们想开得很快，而是他们那时已经有了快感，却根本没有意识到自己是在超速行驶，因此在造成交通事故后大都感到后悔。针对这种状况，开发超速报警器显得尤为重要。如果汽车驾驶员们在各自汽车上安装了汽车超速报警器，在极大程度上能避免发生严重的交通事故。

CMOS High-Speed IOs — Present and Future

CMOS High-Speed I/Os—Present and FutureM.-J.Edward Lee1,William J.Dally1,2,Ramin Farjad-Rad1,Hiok-Tiaq Ng1, Ramesh Senthinathan1,John Edmondson1,and John Poulton11Velio Communications,Inc.2Stanford UniversityMilpitas,CA Stanford,CAAbstractHigh-speed I/O circuits,once used only for PHYs,are now widely used for intra-system signaling as well because of their bandwidth,power,area,and cost advantages.This technology enables chips with over1Tb/s of I/O bandwidth today and over10Tb/s of bandwidth by2010as both sig-naling rates and number of high-speed I/Os increase with process scaling.Key technologies that enable this growth in I/O performance include low-jitter clock circuits and equal-ized signaling.An analysis of clock jitter and channel inter-ference suggests that signaling rates should track transistor performance to rates of at least40Gb/s over boards,back-planes,and short-distance cables.1.IntroductionHigh-speed input/output circuits are becoming increas-ingly critical as technology scales to increase system band-width and decrease power dissipation,die area and system cost.Once used primarily for serial PHYs,high-speed I/O circuits are rapidly becoming the technology of choice for all intra-system connections as well.High-speed I/Os inte-grated in large numbers enable chips with over1Tb/s of I/O bandwidth today.Furthermore,the per-pin bandwidth scales with device speed,at≈20%per year.As this trend continues,chips with many hundreds of20Gb/s I/Os will be feasible by2010.High-speed I/Os use incident-wave signaling in which a signal is detected on itsﬁrst traversal of the signal line(the incident wave)and absorbed by a receive termination.This enables the data bandwidth to scale with transistor perfor-mance,independent of the length of the line.At high data rates,several bits may be in transit at once—pipelined along the length of the line.In contrast,traditional I/O de-signs,e.g.,LVCMOS,have a bandwidth that is limited by the length of the signal line rather than transistor perfor-mance.Without matched terminations,these I/O systems have to ring-up the signal wire over several round-trip de-lays to reliably send one bit.Their data bandwidth is tied to the length of the line,independent of transistor perfor-mance,resulting in a bottleneck as bandwidth demands in-crease.More than half of the power dissipation of many sys-tems today is I/O power,and the fraction of power due to I/O is increasing.The dynamic power of a logic function scales asα3(where gate length scales asα)while a portion of I/O power scales only withα,because a certain amount of current must be delivered to a load that is matched to the line impedance to reliably detect the signal.The mini-mum current required per I/O is nearly constant,indepen-dent of bit rate;thus high-speed I/Os give more bandwidth for thisﬁxed power.Furthermore,the additional power re-quired to build a sophisticated high-speed I/O often scales withα3,like the core logic.Thus,a better process technol-ogy not only enables a higher bandwidth per channel but also reduces the energy consumed per bit.There are two fundamental challenges to continued scal-ing of high-speed I/Os:band-limited channels and timing uncertainty.As data rates increase,channel bandwidth be-comes limited by the frequency-dependent loss(FDL)of the channel.The distance that a signal can be reliably prop-agated decreases with the square-root of signal bandwidth for cables(where skin-effect dominates)and linearly with signal bandwidth for circuit boards(where dielectric ab-sorption dominates).Equalization can cancel the frequency-dependent part of the attenuation.However,the magnitude of the attenuation is ultimately a limiting factor.Also,as at-tenuation levels increase,care must be taken to avoid near-end cross-talk,which is becoming a signiﬁcant problem in legacy systems.As signal rates scale,the timing jitter of a high-speed I/O must decrease to remain a constant fraction of a bit time or unit interval(UI).Power supply noise,substrate noise and thermal noise are the most important contributors to clock jitter.Fortunately,our analyses show that by increas-ing reference clock frequencies and devoting a larger frac-tion of I/O area to clock circuits,timing jitter can be made to scale with bit time.Overall,it appears that there are no major obstacles to achieving40Gb/s signaling rates over boards,backplanes,and short-distance cables(tens of me-ters).Hence signaling rates should continue to scale with transistor performance to at least this speed.The remainder of this paper describes high-speed I/O cir-cuits in more detail.Section2describes the architecture of a typical high-speed I/O and the details of some of its com-ponents.Section3discusses the current state-of-the-art in high-speed I/O technology and the future challenges posed by channel attenuation and clock jitter.2.A Typical High-Speed I/O2.1.Top Level ArchitectureFigure1shows a typical high-speed I/O1.The transmit-ter converts N-bits of parallel data from the core logic into a two-bit stream,and then a2:1multiplexer gates out two symbols per clock cycle with precise timing.The retimer ensures the data are positioned correctly for multiplexing.A higher multiplexing ratio can be implemented with more clock phases to further reduce the frequency requirement. However,in multiplexed systems any phase mismatch be-tween different clock phases results in deterministic jitter in the serialized data.To avoid this,the data can be retimed with a full bit-rate clock before theﬁnal output driver,at the expense of higher power consumption and a lower data rate[2].A pseudo-random bit sequence(PRBS)generator is usually built in for at-speed testing.A bank of samplers in the receiver samples the bit stream on evenly spaced clock phases to demultiplex the data di-rectly,easing the frequency requirement.This multiphase approach suffers the same deterministic jitter problem as its counterpart at the transmitter.The clock recovery unit ad-justs the clock phase to place the data samples in the mid-dle of the bit cell.The adjustment is performed by sampling both the center(the samplers labeled C)and edge(the sam-plers labeled E)of each bit cell.On a transition,the value of the edge sample determines if the sampling clock is early or late2.The2-bit data from the samplers are deserialized into an N-bit parallel data suitable for the digital logics. 2.2.Bandwidth LimitationsThe bandwidth achievable by a signaling system is lim-ited by attenuation,interference,and jitter.These factors are illustrated in the conceptual eye diagram of Figure2.Con-structed by folding the data waveform into a symbol time, an eye diagram shows variations of signal amplitude(volt-age noise)and timing(jitter)across bit cells.The rectan-gle in the middle represents the eye opening,which must be wider than the receiver jitter plus aperture3and taller than 1In this example,a signal can only go in one direction on a channel.Si-multaneous bidirectional signaling allows signals toﬂow in both di-rections on one channel but will not be discussed in this paper since it is rarely encountered.2This type of phase detector,called a bang-bang or Alexander phase detector,is popular due to its ease of implementation[1].the receive sensitivity.That is,t b≥t r+t a+t u,all,where t b is the bit time,t r is the rise time,t a is the receiver aper-ture,and t u,all is the total timing uncertainty of the system [4].For most systems,the dominant component is t u,all, caused mostly by clock jitter,intersymbol interference(ISI) and cross-talk.Input offset is often the largest component of the re-ceiver sensitivity.As shown in Figure3,a digital calibration scheme can cancel this offset with digitally trimmed current sources trained at startup[6][11].With this method,the off-set is reduced from>100mV to<10mV.2.3.Clock MultiplierA clock multiplier multiplies the reference clock up to the multiplexing rate.Two common implementations are a phase-locked loop(PLL),shown in Figure4,and a multi-plying delay-locked loop(MDLL),shown in Figure5.In a PLL,the bandwidth of the feedback loop should be high to reject the oscillator jitter since a low-jitter reference clock is often provided(e.g.,from a crystal).In practice,how-ever,the bandwidth of a PLL is limited to about5%of the reference clock frequency due to the delay around the loop [9][14].In contrast,an MDLL,shown in Figure5,period-ically injects the clean reference clock into the oscillator to reset the phase error every reference clock cycle[5][12]. In this implementation,the pulser generates an enable pulse for the multiplexer(so that the clean reference clock can be muxed in)and the phase detector(so that only one oscilla-tor edge per reference clock cycle is compared).Figure6illustrates the response of a PLL and a MDLL to a frequency shift(or a phase ramp),which is a common test for the jitter performance of clock generation circuits since supply noise exhibits a similar behavior.Jitter in a PLL ac-cumulates until the loop is able to respond.Both the peak jitter amplitude and the time it occurs are approximately in-versely proportional to the loop bandwidth.In contrast,the clean reference clock resets the jitter in a MDLL every ref-erence clock cycle.The peak of the sawtooth decays as the loop gradually corrects the frequency offset.It can be shown that even with the upper bandwidth limit,a PLL exhibits more than twice the peak jitter amplitude compared with a MDLL,which does not require a high loop bandwidth to achieve low jitter.2.4.Clock RecoveryThe clock recovery block determines where to position the sampling clocks.A PLL locked to the receiver input is often used for this function.Unlike the clock multiplier at3Receiver aperture is the time the signal must be above the receiver sen-sitivity to make a correct decision.Data OutFigure1:A typical high-speed I/O architecture.trtu,rxintasensitivityFigure2:A conceptual eye diagram.Figure3:A digitally-trimmed input sampler.the transmitter,the high-jitter receiver input necessitates alow loop bandwidth,which is in conﬂict with oscillator jitterrejection.A dual-loop approach,in which a high-bandwidthclock multiplier is used to multiply a low-jitter referenceclock and a separate low-bandwidth loop is used for receiverinput tracking,removes this tradeoff[19].Most dual-loop systems use aﬁrst-order receiver track-ing loop,as shown in theﬁrst dashed box in Figure7.Thebinary early/late indications from the phase detector arepassed through a phaseﬁlter to reduce noise due to inputjitter.Theﬁlter output controls the phase of the samplingclocks through a timing vernier that changes the phase ofthe clock multiplier output.With a plesiochronous input,this results in either phase lag,if the loop is too slow to trackRefFigure4:A phase-locked loop.RefbckackenidealFigure5:A multiplying delay-locked loop.the input,or phase wander,if the loop is too fast toﬁlter theinput jitter,or both.A second-order receiver tracking loopeliminates these phase errors by estimating the frequency ofthe input signal.The frequency tracking loop,shown in thesecond dashed box in Figure7,integrates the output of thephaseﬁlter to estimate the frequency of the received sig-nal and sends a stream of up/dn signals to compensate forany offset from the reference clock frequency.This enablesa slow loop to be used toﬁlter input jitter without causingphase lag[13].The advantage of using a digital implemen-tation is that many loop parameters,such as the length ofthe phaseﬁlter and the frequencyﬁlter,can be made pro-TimeJitterPLLMDLLT refFigure 6:Jitter Response of a PLL and a MDLL.Rx Figure7:A digital clock recovery architecture.grammable (e.g.,to maximize jitter ﬁltering or minimize lock time).Furthermore,the digital control to the timing vernier can be easily bypassed to allow ﬂexible position-ing of the sampling clocks for testing purposes.2.5.EqualizationSkin effect,dielectric absorption and discontinu-ities cause a channel to exhibit frequency-dependent loss (FDL).A pulse representing a bit not only gets atten-uated by the channel but is spread out in time,caus-ing ISI.Figure 8showsthe frequency response and the 6.25Gb/s pulse response of a backplane channel.A sig-niﬁcant amount of ISI is present at the adjacent sam-ple points in the pulse response (the vertical grid lines are spaced at the sample points,160ps apart).A ﬁlter,or equalizer,with an inverse channel response can be used to counteract FDL.A commonly used ﬁlter is a discrete-time symbol-spaced FIR ﬁlter.Oftentimes it is im-plemented at the transmitter (transmitter pre-emphasis)with direct current summing of different taps at the output [3][11].Figure 8shows the effect of a 4-tap ﬁlter (1main tap and 3post-cursor taps)in the frequency domain and time domain on the same backplane channel.Since a portion of the available transmitter current is assigned to the equaliza-tion taps,in effect transmitter pre-emphasis attenuates the low-frequency component to achieve a ﬂat spectrum over-all.With pre-emphasis,the amount of ISI is signiﬁcantlyFigure 8:Response of a backplane channel.in+in-Figure 9:Active RC-based receive equalization.reduced.As shown in Figure 8,it opens up a completely closed eye (PRBS 23pattern).Sometimes it is beneﬁcial to place the equalizer at the re-ceiver.Although a discrete-time FIR approach can be used,it is signiﬁcantly more complicated than transmitter pre-emphasis since high-speed sampling,multiplication,and addition of analog values are required.An alternative is an active high-pass ﬁlter,shown in Figure 9[6].The gain of this circuit goes up with frequency as the capacitor de-creases the amount of source degeneration.The equaliza-tion gain can be adjusted through the variable resistors.3.Future ChallengesAs gate lengths are scaled by α(at a rate of about 20%per year),gate delay also scales as αand transistor ωT scales as 1/α.Signaling bandwidth can also scale as 1/αif the timing uncertainties,dominated by clock jitter and chan-nel interference can be made to scale at the same rate.This section investigates the scalability of clock jitter and dis-cusses how channel interference can be improved throughcircuit level and system level techniques.With careful cir-cuit and system design,we expect the bandwidth of elec-trical signals on boards,over backplanes,and over cablesto scale to at least40Gb/s.I/O energy per bit is expected to scale asαtoα2in the near future,but will eventuallybe limited byα.In contrast,the switching energy per func-tion for digital logic scales asα3.As a result,the fraction ofI/O power in a system will increase for the foreseeable fu-ture.3.1.Scalability of Clock JitterAnalysis of a CMOS inverter ring oscillator suggests that clock jitter can be made to scale withαif higher refer-ence clock frequencies are used and if an increasing per-centage of I/O area and power is devoted to clock genera-tion.We investigate the effects of the three most importantnoise sources:power supply noise,substrate noise,and ther-mal noise.3.1.1.Power Supply Noise A k%change in supply volt-age results in a k%change in the period of a CMOS ring oscillator.Assuming that supply noise remains a constantfraction of the supply,if the reference clock frequency re-mains constant,the p-p jitter will remain constant since boththe rate and the duration of jitter accumulation areﬁxed.In other words,jitter as a percentage of the bit time in-creases.To ameliorate this problem,we can increase thesupply noise rejection and/or increase the reference clockfrequency.Local supply regulation,shown in Figure10,is com-monly used to isolate critical circuits[11].On-chip digi-tal switching often generates signiﬁcant supply noise.Toﬁrst-order approximation,the amount of noise rejection bythis type of regulator is proportional to C1/C2.Therefore, supply rejection can always be improved with area.It alsoimproves with process scaling as long as the area of C1 scales slower thanα2.For multiphase oscillators,however, the area of the delay element often needs to remain constant to keep phase mismatch aﬁxed fraction of the bit time.In this case,the area of C1must increase with process scal-ing to improve the supply rejection.A bit-rate oscillator is advantageous in this regard since it does not rely on match-ing of the delay stages to produce precise clock phases.The frequency of the crystal reference is limited by its thickness and cannot be expected to scale as aggressively as the semiconductor technology.Since on-chip LC oscil-lators exhibit a much better jitter performance,it is advan-tageous to multiply the reference clock to an intermediate frequency with a global on-chip LC oscillator and use local ring oscillators to generate theﬁnal high-frequency clocks whenever integration or tunability is a concern.Fortunately,VregFigure10:A local supply regulator.the Q of on-chip inductors is improving with the availabil-ity of more metal layers in advanced CMOS processes.With a combination of higher reference clock frequen-cies and better supply noise rejection,jitter induced by power supply noise should continue to scale with the bit time in the foreseeable future.3.1.2.Substrate Noise,also caused mostly by digi-tal switching,is a major concern in highly integrated applications.Fortunately,process remedies are now read-ily available to reduce its effect.For example,many processes now offer deep NWELL to isolate a subcir-cuit from the rest of the chip.Recent work has demon-strated better than50dB attenuation of substrate noise with only200µm of separation in an epi process[8].Ju-dicious use of this structure should keep substrate noise a negligible effect on sensitive circuits such as clock genera-tors.3.1.3.Thermal Noise Unlike supply and substrate noise whose magnitude can be attenuated externally,ther-mal noise is inherent in the device4.The rms jitter of an N-stage CMOS ring oscillator when placed in a PLL or a MDLL is[10][15]σjit=Γrms2πf0CV swN2i2n∆fτL(1)where f0is the oscillator frequency.For a PLL,τL is 1/2πf L,where f L is the loop bandwidth.For a MDLL,τL is1/f ref,where f ref is the reference clock frequency.Γis the impulse sensitivity function(ISF)and determines the sensitivity of the oscillator to a noise impulse[10].For ex-ample,noise occurring at the edge of the clock produces more jitter than that at the peaks.It can be shown thatΓrms scales asα1.5due to sharper edges at higher frequencies. CV sw is the maximum charge swing and determines how easily the oscillator nodes can be moved.It scales asα2. i2n/∆f is the amount of thermal noise on one node and re-mains approximately the same with scaling5.This analysis4Although MOSFETs suffer from poor1/f noise,its effect on jitter can be reduced signiﬁcantly by balancing the rise and fall time of the clock.Furthermore,low-frequency jitter such as1/f noise is attenu-ated when placed in a high-bandwidth loop.indicates that while the clock period scales as α,the rms jit-ter scales as √αfor a ﬁxed reference clock frequency and as αif the reference clock frequency scales at the same time.Furthermore,increasing the width of the delay element im-proves jitter in a square root fashion due to a higher charge swing.It is instructive to compare the magnitude of jitter in-duced by thermal noise and that induced by supply noise.Recent measurement of a 0.25µm 1.33GHz CMOS ring oscillator showed a thermal-noise-induced phase noise of -111.5dBc/Hz at a 1MHz offset from the carrier [10].For a MDLL with a multiplication factor of 10,this roughly translates into a rms jitter of 0.173ps at the end of a ref-erence clock period.The p-p jitter for <10−15probabil-ity is 2.77ps.In contrast,a 5%supply noise with a 20dB power supply rejection results in roughly 37ps p-p jitter.In summary,by increasing the reference clock frequency and increasing the oscillator width,thermal-noise-induced jitter should scale well with the bit time.In addition,in highly integrated applications,thermal noise will likely re-main a negligible effect for the foreseeable future.3.2.ChannelHigh-speed I/Os are typically used between chips on a printed circuit board,across a connectorized backplane,and across short distance cables (tens of meters).The FDL (in dB)scales linearly with bandwidth for typical circuit boards,where dielectric absorption dominates,and as the square-root of bandwidth for cables,where skin effect dom-inates.In addition,discontinuities can cause signiﬁcant FDL beyond these fundamental loss mechanisms.While equalization can ﬂatten the spectrum of these channels,to-tal attenuation as well as external interferences will ulti-mately limit the achievable bit rate.In this section we fo-cus on backplane channels because they are the most chal-lenging in terms of attenuation and cross-talk.For many systems (e.g.,switches and routers),band-width is upgraded through gradual replacement of cards in an existing backplane.These legacy backplanes,suitable for the speed requirement at the time they are designed,often exhibit very high attenuation and very low signal-to-interference ratio (SIR)as bit rate increases.Figure 11shows the cumulative distribution functions (CDFs)of ISI,8cross-talk aggressors,and the total (ISI plus cross-talk)for one such backplane (same channel as Figure 8)running at 6.25Gb/s.The right side of the plot stops at the amplitude of the received pulse.Therefore,the probability of a bit er-ror due to a particular interference is the intersection of the5It has been observed that thermal noise increases in deep-submicron technologies due to high-ﬁeld carrier heating.Fortunately,voltage scaling helps reduce this effect.P r o b a b i l i t y Figure 11:CDFs of channel interferences for a backplane,including the effects of clock random jitter (RJ)and deter-ministic jitter (DJ).curve with the y-axis.Although a 4-tap equalizer is used,the bit-error-rate (BER)is still unacceptably high at 10−7.The decreasing signal-to-inteference ratio is best man-aged through a combination of circuit level and system level improvements.Currently,most high-speed I/Os use a 2-tap linear ﬁlter that is manually adjusted through either trial-and-error or channel analysis.As longer ﬁlters are required to further remove the ISI,adaptive equalization ,in which the tap coefﬁcients are optimized by hardware,becomes an a critical requirement [20].It not only obviates the need for user intervention that is often time-consuming but also im-proves the effectiveness of equalization by including the effects of package and termination non-idealities that are lost in s-parameter or eye measurements.A non-linear ﬁl-ter,such as a decision-feedback equalizer (DFE),can fur-ther improve the margin by equalizing the signal without amplifying the cross-talk.In contrast,a high-pass linear ﬁl-ter commonly used to equalize the channel ampliﬁes the high-pass cross-talk signiﬁcantly.Because the channel response attenuates while the cross-talk ampliﬁes at high-frequencies,sending more bits per unit bandwidth through multi-level signaling is an attractive way to manage this problem [7].Figure 12compares binary and 4-level eye diagrams for the same symbol rate .The hor-izontal eye opening of 4-level signaling is less than binary signaling due to limited slew rate.Furthermore,its vertical eye opening is less than 1/3that of binary signaling due to the voltage noise at the intermediate level (V n ).Multi-level signaling often requires additional overhead bandwidth to ensure enough useful transitions 6are present for clock re-covery.The exact beneﬁt of multi-level signaling needs tobe simulated on a per-channel basis,performing an anal-ysis similar to that shown in Figure11.However,a useful rule-of-thumb is that the SIR must increase by at least12dB in the octave from1/4to1/2the bit rate for4-level signal-ing to be advantageous.Careful system design is needed in addition to circuit level innovations to sustain continuous bandwidth scaling.For example,via stubs often cause FDL to be much worse than expected from skin effect and dielectric loss due to quarter wavelength resonance.Back-drilling,in which theunused portion of the via is removed,provides a very cost-effective way to push out this resonance[18].Without back-drilling,a180mil thick FR4backplane via stub creates a resonance at about5GHz for typical via sizes.The primary source of cross-talk in most systems is the backplane connector.New connectors are being introducedwith ground shields completely surrounding each signal pair to reduce cross-talk.Signalsﬂowing in opposite direc-tions are isolated from each other to avoid near-end cross-talk,which is much more detrimental than far-end cross-talk since the interference is not attenuated by the full length ofthe channel along with the signal.Cross-talk coupling less than-50dB has been demonstrated on a typical backplane with these improvements[18][17].With a50mV receiver sensitivity now available in com-mercial high-speed I/Os,26dB of FDL at1/2bit rate can be tolerated for a typical1V p-p ing the tech-niques mentioned above,along with low-loss laminates, <20dB of FDL up to10GHz has been demonstrated on fully connectorized backplane channels up to70cm.10Gb/sdata transmission without any equalization has been demon-strated,and20Gb/s data transmission with simple2-tap pre-emphasis is now possible[17][16].With further in-vestment,it appears that achieving<30dB FDL up to 20GHz and one meter is not out of reach.This,combined with further process and circuit improvements on receiver sensitivity,jitter,and equalization,should enable a40Gb/s transceiver over backplanes in the future.Of course,these beneﬁts cannot be fully realized unless the whole system, including the backplane,is completely upgraded.3.3.Current State-of-the-Art and Future TrendFigure13shows that the bandwidth of production back-plane channels has been doubling every two years since 1999.3.125Gb/s channels are now commonplace and 6.25Gb/s and10Gb/s channels have been demon-strated[6][21].It is clear that this bandwidth growth trend is not sustainable since device speed is only dou-6Only transitions from outer levels to outer levels and inner levels to inner levels are useful for clock recovery.Figure12:A comparison of binary and4-level eye dia-grams.bling every3–4years.Techniques such as multi-level sig-naling only provide a one-time bandwidth increase.Since 1999,I/O technology has been catching up to the semicon-ductor technology,making the super-Moore’s-Law band-width trend possible.A practical limit of the symbol timefor high-speed I/Os is about2FO4(fanout of4inverter de-lay).In0.13µm CMOS technologies,this limit is about 7Gb/s(or12Gb/s for4-level signaling).It is expected thatthe per-channel backplane bandwidth growth will be lim-ited by semiconductor scaling beyond10Gb/s and at least up to40Gb/s when the channel imperfections be-come the critical bottleneck.High-speed I/O energy per bit will ultimately be lim-ited by the transmitter output drive,which requires at least a constant current to overcomeﬁxed noise and higher lossin the channel.As a result,transmit energy per bit scales as α.For a CMOS inverter based multi-phase clock multiplier, energy per bit also scales asαsince gate capacitance mustincrease to scale transistor mismatch.For a bit-rate oscilla-tor,where matching is less of an issue,energy per bit scales asα3.The rest of the circuits,including the transceiver data paths and the digital clock recovery unit,are digital logic and hence scale asα3.As a result,the energy per bit for one high-speed I/O is expected to scale asαtoα2in the near future,but will eventually be limited byα.In compar-ison,the switching energy for a digital logic function scales asα3.Thisα2difference in scaling is partly offset by in-creased integration.As G=1/α3more core logic band-width is integrated on a chip(holding total core power con-stant),Rent’s rule suggests that only I=G2/3=1/α2 more I/O bandwidth will be required,consuming1/αtimes as much I/O power.The ratio of I/O power to core power on a chip will hence increase by1/αwith technology scal-ing.4.ConclusionHigh-speed I/O circuits enable signaling rates to scale with device technology,largely independent of transmission。

毕业设计--超速报警显示系统设计与制作

1.2 数字式转速测量系统的发展背景
目前国内外测量转速的方法很多，按照不同的理论方法，先后产生过模拟测速法(如离心式转速表、用电机转矩或者电机电枢电动势计算所得)、同步测速法(如机械式或闪光式频闪测速仪)以及计数测速法。计数测速法又可分为机械式定时计数法和电子式定时计数法。传统的电机转速检测多采用测速发电机或光电数字脉冲编码器，也有采用电磁式(利用电磁感应原理或可变磁阻的霍尔元件等)、电容式(对高频振荡进行幅值调制或频率调制)等，还有一些特殊的测速器是利用置于旋转体内的放射性材料来发生脉冲信号。而霍尔传感器具有非接触、长寿命、低噪音、高精度、不受震动和灰尘影响等优点，特别适合于测量汽车转动部件的转速。
Keywords: speed；SCM；sensor
第一章
1.1 本设计课题的目的和意义
随着汽车工业的不断进步,行驶在道路上的车辆越来越多,交通事故发生的频率也不断增加。众所周知，交通事故的发生大部分是由驾驶员的超速驾驶造成的。为提高汽车运行的安全性，减少交通事故的发生，本文讲述了一个由单片机控制的车速报警系统。
P1口：P1口是一个内部提供上拉电阻的8位双向I/O口，P1口缓冲器能接收输出4TTL门电流。P1口管脚写入1后，被内部上拉为高，可用作输入，P1口被外部下拉为低电平时，将输出电流，这是由于内部上拉的缘故。在FLASH编程和校验时，P1口作为第八位地址接收。
P2口：P2口为一个内部上拉电阻的8位双向I/O口，P2口缓冲器可接收，输出4个TTL门电流，当P2口被写“1”时，其管脚被内部上拉电阻拉高，且作为输入。并因此作为输入时，P2口的管脚被外部拉低，将输出电流。这是由于内部上拉的缘故。P2口当用于外部程序存储器或16位地址外部数据存储器进行存取时，P2口输出地址的高八位。在给出地址“1”时，它利用内部上拉优势，当对外部八位地址数据存储器进行读写时，P2口输出其特殊功能寄存器的内容。P2口在FLASH编程和校验时接收高八位地址信号和控制信号。

公路车辆高清智能监测报警系统建设方案

【公路车辆高清智能监测报警系统】建设方案XXXXXX有限公司目录一. 概述 ............................................................................................................................................. - 4 -1.1 引言 .................................................................................................................................. - 4 -1.1.1 项目背景 ...................................................................................................................... - 4 -1.1.2 设计目标 ...................................................................................................................... - 5 -1.1.3 系统设计的指导思想................................................................................................ - 5 -1.1.4 系统设计的基本原则................................................................................................ - 6 -1.1.5 设计、制造及安装标准依据.................................................................................. - 7 -1.2 产品介绍 .................................................................................................................................. - 9 -二.公路车辆高清智能监测报警系统 ............................................................................................ - 10 -2.1 系统结构图........................................................................................................................... - 10 -2.2 系统布局 ............................................................................................................................... - 11 -2.2.1双向四车道布局示意图......................................................................................... - 11 -2.2.2前端车辆通行感知单元......................................................................................... - 12 -2.2.3图象传输单元 ........................................................................................................... - 12 -2.2.4中心数据管理单元 .................................................................................................. - 14 -2.3系统原理 ................................................................................................................................ - 15 -2.4系统特点 ................................................................................................................................ - 16 -2.5 系统功能 ............................................................................................................................... - 17 -2.5.1 车辆捕获 ................................................................................................................... - 17 -2.5.2 图像存储 ................................................................................................................... - 18 -2.5.3 车辆牌照识别 .......................................................................................................... - 18 -2.5.4 车辆测速 ................................................................................................................... - 19 -2.5.5 联网布控 ................................................................................................................... - 19 -2.5.6 流量统计 ................................................................................................................... - 19 -2.5.7 套牌车辆比对 .......................................................................................................... - 20 -2.5.8 嫌疑及违法车辆拦截............................................................................................. - 20 -2.5.9 数据信息实时处理 ................................................................................................. - 20 -2.5.10 系统防雷 ................................................................................................................. - 21 -2.5.11 软件检测报警参数可设置 ................................................................................. - 21 -2.5.12警务终端 .................................................................................................................. - 21 -2.5.13卡口监控 .................................................................................................................. - 21 -2.5.14 故障自动检测及恢复 .......................................................................................... - 21 -2.5.15 系统抓拍效果 ........................................................................................................ - 22 -三.系统设备预算清单........................................................................................................................ - 24 -一. 概述1.1 引言1.1.1 项目背景现代交通管理领域，高速公路和主要干道的建设水平不断提高，道路环境正在逐步改善。

高速公路超速预警系统设计

高速公路超速预警系统设计高速公路是现代化快速交通方式，随着交通工具速度的提升，高速公路安全问题越来越引起关注。

超速是高速公路上发生事故的主要原因之一。

因此，高速公路超速预警系统的设计就变得非常重要了。

高速公路超速预警系统是一种利用现代计算机技术和通信技术进行处理和监测的预警系统，它主要是通过路侧设备、车载终端和中心服务器等来实现的。

它能够及时发现车辆超速情况，并进行及时的警示，从而提醒驾驶员注意安全驾驶，防止交通事故的发生。

高速公路超速预警系统主要由以下几个方面组成：1.路侧设备：路侧设备是超速预警系统的重要组成部分，它通过远程监测车辆的车速等信息，当车辆超速时，路侧设备会向车载终端发送警示信息，提醒驾驶员减速。

2.车载终端：车载终端是超速预警系统的另一个重要组成部分，它主要是监测车辆的车速等信息，并将这些信息传递给中心服务器。

当车辆超速时，车载终端会通过声音、振动和显示屏等方式向驾驶员发出提示。

3.中心服务器：中心服务器是超速预警系统的核心组成部分，它主要是用于接收和处理车载终端传递的信息，并向路侧设备和车载终端发送指令。

同时，中心服务器还可以对数据进行分析和统计，帮助交通管理部门进行交通流量分析和路况调度。

高速公路超速预警系统的工作流程如下：1.路侧设备实时监测车辆的速度等信息，将数据上传到中心服务器。

2.中心服务器接收到车辆信息后，通过计算与数据库的比对，判断车辆是否超速。

3.如果车辆超速，中心服务器向路侧设备发送警示信息，并将信息发送到车载终端。

4.车载终端接收到信息后，通过声音、振动和显示屏等方式向驾驶员发出提示，并采取措施减速。

在设计高速公路超速预警系统时需要考虑以下几个方面：1.可靠性：超速预警系统对于驾驶员的生命安全是至关重要的，因此它必须具有高可靠性。

路侧设备要具有防水、防腐、耐高温、耐低温等特性，设计合理的电力控制保护装置。

车载终端需要选择可靠耐用、功能强大的智能终端。

2.实时性：超速预警系统需要时刻进行实时监测，及时发现车辆超速情况，因此它必须具有很高的实时性。

运输超长超宽货物大型车辆右转实时告警系统设计

运输超长超宽货物大型车辆右转实时告警系统设计发布时间：2022-05-13T03:13:03.838Z 来源：《福光技术》2022年10期作者：陈兴林杨伟苓陈毛[导读] 随着现代工业的发展，超长超宽的大型设备及大型构件（如图1所示风电叶片、压力容器等）越来越多，对现代运输车辆及道路通行条件及安全要求越来越高。

临沂大学自动化与电气工程学院临沂 276600摘要：为弥补运输超长超宽货物大型车辆右转盲区预警难、预警准确效果差等不足，设计了一种可以适应全时段的运输超长超宽货物大型车辆右转实时告警系统。

当转向传感器感知右转向信息时，全系统开始工作，CMOS相机监视右侧盲区，毫米波雷达探测右侧行驶车辆的实时距离与速度，通过Jetson-Tx2处理器进行实时信息处理，依据安全距离判断模型，若判断到危险情况，立即通过蜂鸣器和LED闪烁灯告警。

该系统结构简单、工作原理可靠，为减少运输超长超宽货物大型车辆发生或引起交通事故提供技术支撑。

关键词：汽车工程；CMOS相机；毫米波雷达；危险判据中图分类号：U471.15 文献标志码：A随着现代工业的发展，超长超宽的大型设备及大型构件（如图1所示风电叶片、压力容器等）越来越多，对现代运输车辆及道路通行条件及安全要求越来越高。

这些超长超宽的大型设备及大型构件需办理特殊运输许可证方可上路运输，但是载货超长超宽的车辆上路行驶时存在极大的安全隐患，不仅会严重影响后方车辆的视线，也会影响自身车辆转向的视野。

同时车辆自身重心发生偏移导致车辆在变向、转弯、刹车等紧急情况时易发生侧翻。

尤其右转向时载货超长超宽的大型车辆右侧盲区扩大和内轮差的存在，驾驶员在右转弯时的视野会受到更大的限制，容易发生意外事故。

运输超长超宽货物大型车辆右转实时告警系统是在车辆右侧安装毫米雷达、视觉等传感器，通过对盲区内人的识别和车辆的识别，对盲区存在的危险状况进行预警，既提示驾驶员又警示周围行人与车辆。

运输超长超宽货物大型车辆右转实时告警系统作为一种重要的辅助驾驶系统越来越受到汽车厂商的重视。

园区车辆测速系统解决方案

园区车辆测速系统解决方案园区车辆测速系统解决方案文档控制目录第1章系统方案概述1.1 应用背景随着园区内车辆数量的增加，安全问题日益凸显。

园区管理方需要一种能够监控车辆速度并及时预警的系统，以保障园区内交通安全。

1.2 业务现状目前，园区内的车辆测速主要依靠人工测速，存在测量不准确、工作量大、效率低等问题。

同时，无法及时发现超速行驶的车辆，存在安全隐患。

1.3 需求分析1.3.1 业务需求园区车辆测速系统需要具备以下功能：1）实时监测车辆速度；2）自动识别车牌号码；3）记录车辆违规信息；4）及时预警超速行驶车辆。

第2章系统设计2.1 系统框架园区车辆测速系统分为前端监测设备、中间处理服务器和后端数据管理系统三个部分。

前端监测设备包括摄像头和测速仪器，用于实时监测车辆速度和识别车牌号码。

中间处理服务器用于处理前端传输的数据，记录车辆违规信息。

后端数据管理系统用于存储和管理数据，并进行超速行驶车辆的预警。

2.2 系统流程系统流程包括车辆监测、数据处理和信息管理三个环节。

当车辆通过监测设备时，设备会自动识别车牌号码并记录车辆速度。

数据会传输到中间处理服务器进行处理，判断车辆是否违规行驶。

如果车辆超速行驶，系统会自动进行预警并记录车辆信息。

第3章系统实现3.1 前端监测设备前端监测设备主要包括摄像头和测速仪器。

摄像头用于拍摄车辆照片和识别车牌号码，测速仪器用于测量车辆速度。

设备需要具备高清晰度、高识别率、高测量精度等特点。

3.2 中间处理服务器中间处理服务器需要具备高性能、高稳定性、高可靠性等特点。

服务器需要能够实时处理大量数据，并能够进行数据备份和恢复。

3.3 后端数据管理系统后端数据管理系统需要具备高可扩展性、高安全性、高性能等特点。

系统需要能够存储大量数据，并能够进行数据备份和恢复。

同时，系统需要具备权限管理、数据分析等功能。

结论园区车辆测速系统能够有效解决园区内车辆超速行驶问题，提高交通安全水平。

电子车速里程表的单片机实现方案

电子车速里程表的单片机实现方案清晨的阳光透过窗帘，洒在键盘上，手指轻轻敲击，方案的大致轮廓在脑海中逐渐清晰。

10年的方案写作经验，让我对这类项目有了更深的理解和把握。

咱们就聊聊这个电子车速里程表的单片机实现方案。

这个方案的核心是单片机。

想象一下，单片机就像是一个微型的大脑，控制着整个电子车速里程表的工作。

我们选用的是ST公司的一款高性能、低功耗的单片机，具备丰富的外设接口，足以应对这个项目的需求。

一、硬件设计1.车速传感器车速传感器是整个系统的输入部分，它通过检测汽车车轮的转速，将车速信号传输给单片机。

我们采用的是霍尔效应传感器，具有响应速度快、精度高的特点。

2.里程计数器里程计数器负责记录汽车行驶的总里程数。

这里我们采用了一个32位的计数器，足以满足大多数汽车的使用需求。

3.显示模块显示模块是整个系统的输出部分，负责将车速、里程等信息显示给驾驶员。

我们选用的是一块高亮度的LCD显示屏，清晰度足够，即使在阳光直射下也能看得清楚。

4.电源模块电源模块为整个系统提供稳定的电源，保证系统的正常运行。

考虑到汽车电源的特殊性，我们采用了稳压电路，确保单片机和其他模块在稳定的电压下工作。

二、软件设计1.主程序框架（1）初始化：设置单片机的时钟、IO口、中断等。

（2）车速计算：根据车速传感器的输入信号，计算出汽车的速度。

（3）里程计数：实时更新汽车行驶的总里程数。

（4）显示更新：将车速、里程等信息显示在LCD屏幕上。

2.中断处理（1）车速传感器中断：当车速传感器检测到车轮转速变化时，触发中断，进行车速计算。

（2）按键中断：当驾驶员按下按键时，触发中断，进行相应的操作，如复位里程表、切换显示模式等。

三、系统调试与优化在硬件和软件设计完成后，需要进行系统调试和优化，确保系统在实际运行中的稳定性和可靠性。

1.硬件调试：检查各个模块的连接是否正确，确保电源稳定，传感器信号准确。

2.软件调试：通过模拟各种情况，检查程序的稳定性和可靠性，如车速突变、按键操作等。

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重庆绕城高速公路车速预警系统关键词：高速公路车速预警高速公路车速预警系统是根据重庆市高速公路的特征与高速公路管理服务的要求，组织科技人员进行科技攻关，大胆采用新技术新工艺，于2008年12月26日重庆外环高速公路西段正式通车前成功实施完成的项目。

项目的成功实施，不但为广大行车人员提供了更多、更新、更人性化的安全预警措施，也为提高道路的畅通能力提供了良好的手段，创造了较好的社会效益，得到了重庆市广大市民的好评。

一、项目背景重庆外环高速公路全长１８７公里，共分为东段、北段、南段、西段四个项目。

外环高速公路是交通运输部确定的重庆地区首条"典型示范路"和全国４条"科技示范路"之一。

由于重庆是典型的山区地带，具有多隧桥、多雨雾的特点，而且高速公路一般车速较快，这些对行车安全存在着严重影响。

为了体现“科技示范路”的需求，业主精心求证，大胆创新，采用了更新的高科技技术，建设了高速公路条件下的车速预警信息提示系统，对于车速过快的车进行预警，保障广大驾驶员的生命财产安全，对于车速过慢，长期占用车道，影响道路行驶速度的车辆也进行预警，提醒行驶车辆进行相应处理，从而提高高速公路的道路通过效率和控制好行车速度。

在业主指导下，设计研发集成单位密切配合，克服了众多技术工程难题，创新使用了高速车牌识别技术与视频测速技术，准时完成了系统的建设，为这条崭新科技高速公路的开通提供了高质量的服务，得到了广大行车司机与新闻好评。

高速公路车速预警系统的主要功能是：设置双龙门架，两个龙门架相距300-500米，在第一个龙门架上安装高速车牌照识别视频测速设备，自动完成对经过该断面的所有车辆的车牌识别和车速计算；第二个龙门架上安装有LED显示屏，对经过第一个龙门架的车，一旦车速超出速度设定上下范围时，及时在前方的LED屏上显示相应的车牌号与速度信息，并提醒该车超速或者行驶过慢，司机就能及时清晰看到提示信息，从而减速行驶或者加速行驶，实例示意图如下：二、系统构成系统由以下几个部分构成：外场视频图像采集、外场到机房的传输、车牌识别与视频测速设备、后端数据保存与控制部分的管理主机、外场的信息显示。

外场图像采集部分与外场LED显示部分通常安装在两个相距300-500米的龙门架上。

系统构成如下图所示：①外场图像采集前端图像采集前端包括：高清摄像机、高清镜头、LED辅助光源及辅助光源控制板、室外防护罩。

图像采集前端通常安装在龙门架或L杆上。

外场摄像机的数据通过网络光端机，将IP图片流传送到就近收费站的监控机房，监控机房配置“车牌识别与视频测速处理平台”。

下图为安装在情报板上的一套图像采集前端。

包含3个补光灯（覆盖3个车道），一套内置300万像素CMOS摄像机高清摄像机与镜头的护罩。

安装在防护罩内的高清摄像机和高清镜头②车辆识别平台（车牌识别以及视频测速）车牌识别与视频测速，采用了一套高清视频流检测设备，高清视频流检测设备通过网络口，直接接收前端图像采集设备通过网络光端机传送过来的IP图像流，设备通过动态的IP图片流，在车辆的运动过程中，自动完成车牌识别与视频测速两项功能。

由于图像采集前端设有专门的夜间补光设备，夜间的车牌识别与测速不受影响，可保证全天候24小时不间断的工作。

单台300万像素摄像机可完整覆盖三个车道，抓拍率及识别率高于传统的多台普通摄像机交叉覆盖方案。

正常天气和车辆环境下，该设备输出速度精度：0 到6%；车牌识别率：≥92%。

③控制工作主机一台电脑主机，用于接收高清视频流检测设备的输出信息，分析车辆速度数据，按照设定的速度阀门值（超速或超低速）并能控制外场LED显示屏的显示内容，并根据需要存储车辆信息和对应图片。

④外场信息显示LED显示屏，安装在后一个龙门架上，用于显示超速的车辆车牌与车速，提醒行驶安全信息。

在没有车辆违规信息显示时，LED显示屏显示路段的服务信息。

三、工作原理一、设备连接图像视频采集前端对通过视频监控范围内的目标汽车（一台或者多台）进行逐帧视频检测，识别视频图像范围内的所有车辆的车牌号码并输出对应车牌号码的车速，通过管理主机收集数据，分析车辆信息，输出超速车辆的行驶信息，通过LED显示屏显示，提示驾驶员减缓车速，保障安全行驶。

二、数据流程高清识别器通过对高清摄像机提供的逐帧视频信息进行相关计算，输出车辆行驶的车道信息、车牌号码、车牌颜色和车速，数据同时上传给管理主机，由管理主机根据车辆的速度与车牌号码，输出给LED显示屏加以显示。

三、视频测速原理标定摄像机的垂直高度和与车道的水平位置，标定摄像机的视野中的远端距离及近段距离，在平直的车道上车牌识别器即可实现视频测速。

车牌识别器是逐帧实时处理的，可以画出车辆行驶在摄像机视野范围内的精确轨迹以及记录车辆在所有参考点时刻，通过计算分析，获取车辆的行使速度。

设备提供设置软件，通过该软件的设置，如下图所示，设备就可以实时输出每一辆经过的车辆的车牌号与该车的行驶速度。

四、管理主机软件功能管理主机是系统的核心组成部分，用于前端设备的数据采集、数据存储、实时计算、参数配置管理、驱动显示等功能，详细说明如下：✧接收车牌识别设备实时传送的数据（车牌号、颜色、车道、车速等信息）✧保存车牌识别设备传送过来的所有数据，便于今后分析✧根据识别信息计算出该车是否需要在LED屏上进行相应提醒，每个车道的速度阀门值在配置中设定一个范围，对于超出设定范围的过快或者过慢的车辆，都可以进行预警提醒✧配置功能，可针对每个车道不同颜色车牌配置不同的速度阀门值，包括某类颜色的车牌最低通过速度与最高通过速度值✧合成车牌号码与提醒词，驱动LED屏显示五、项目实际外场图现场图一：视频采集前端✧龙门架高度通常6.6米高，双龙门架模式，在前龙门架的中央设置前端视频图像采集设备，后龙门设置相应的LED显示屏。

✧LED显示屏可以显示行驶车辆的信息，以提醒驾驶员减慢车速。

现场图二：车辆经过前龙门架后，300米处就能够清楚看到自己的车牌与预警提示信息，及时调整行驶速度。

六、核心设备指标摄像机的性能指标➢图像传感器：1/2”CMOS➢彩色最大分辨率：300万象素（2048 X 1536）➢灵敏度：***********➢动态范围：60dB➢最大信噪比：45dB➢最大数据传输速度：45Mbps➢最大视频传输帧数：15fps，2048 X 1536➢图像压缩：22级品质的M-JPEG➢传输协议：TFTP➢网络接口：100Base-TX以太网接口➢外形尺寸：76W X 63.5H X 125L（mm）（不含镜头）➢镜头接口：C/CS镜头卡口➢供电方式：以太网供电（POE）或直流输入（9～12VDC）（2.1mm×5.5mm,中心正极）➢最大功耗：4W镜头的性能指标➢捕捉兆级像素照相机的全部分辨率➢低变形率(低于1.0%)➢紧凑式设计(直径：35.5mm)➢焦距：25mm➢镜头直径于焦距直径的最大值：1:1.4➢图像最大尺寸：8.8mm X 6.6mm（Ø 11mm）➢光圈：手动➢聚焦：手动➢后焦点：13.1mm➢固定架：C固定架高亮辅助光源➢外壳材质：防水铝壳➢光通量：1920流明➢照射距离：0.5～70米➢额定功率：35W➢供电方式：AC 220V/50Hz ±10%➢开关控制电压：0V（关闭），10V（开启）➢净重：5Kg（不包括云台）车牌自动高清识别器➢识别符合“GA36-92”（92式牌照）和“GA36.1-2001”（02式牌照）标准的民用车牌照和97式、04式新军车牌照与07式新武警车牌照的汉字、字母、数字、颜色等信息➢支持标准双层牌识别➢全天候连续工作，适应白天、黑夜、雨雪天气环境➢触发方式类型：无触发➢CPU性能：一秒钟处理25帧全图图像的车牌识别➢单车牌识别时间：＜0.4秒➢整体识别率：＞85%➢车辆检测率：＞95%，彩色图像，可以实现JPEG格式，不小于16位➢输出信息：车辆大图、车牌小图、二值化图、车牌号码和车牌颜色，车辆速度➢视频速度误差：0- -6％（速度超过100KM/H）或者0- -6公里（速度低于100KM/H）➢机箱防护等级：IP50➢工作环境温度：-20℃～+85℃（处理单元），工作湿度：< 95%➢平均无故障时间：MTBF ≥30000小时➢频闪光源寿命：≥20000小时➢允许车辆行驶速度：0 ~ 220km/h➢输出大图的图像格式：1360*1024（140万像素格式）、1600*1200（200万像素格式）、2048*1536（300万像素格式）和2560*1920（500万像素格式）➢设备通讯接口：以太网接口RJ45，10M/100M/1.0G自适应➢视频输入方式：以太网接入➢整机功耗：＜180W➢设备尺寸：430 ×260 ×44.5mm（长*宽*高）➢产品尺寸：430 X 260 X 44.5mm（长*宽*高）➢供电方式：AC 220V/50Hz ±10%七、项目总结车速预警系统应用在复杂高速路段上，预先提醒行驶的车辆，对于车速过快的车进行预警，保障行车人员的生命财产安全；对于车速过慢，长期占用超车道，影响道路行驶速度的车辆也进行预警，提高道路的通行速度。

系统的建设成功，改善了高速公路的服务形象，提高了高速公路的服务质量，具有的社会效益和经济效益。

项目建设中采用了最新的技术及设备，不但使系统结构大大简化，节省了项目建设成本，减少了设备数量，减少了潜在的故障点与维护工作。

实现一体化的车速、车牌、车道、实时显示。

相比较于传统全覆盖多台标清摄像机与雷达等构成的方案，直接采用一套300万像素视频流的车牌识别与测速设备3个车道，系统成本更具性价比，也是整个项目的亮点之处。

方案特点✧采用了一套设备，完成了高速车牌识别与视频测速两项功能，与设备数量少，故障点少，维护工作少，成本少。

✧对于国内大部分2＋1高速主干到，只要采用一套高清前端图像采集设备就可以完整覆盖一条道，没有盲区✧对越线、跨线、并线、S形行驶车辆识别产生的问题，采用高清方案，都能较好处理，后台的不需要运行专门的剔重、查重软件，剔除多检、多牌问题✧高清设备的识别率比多台标清车牌识别设备高✧视频图片场景大，不但可以用于识别，也完全可以替代全景相机的功能六、配置清单1工作量清单（按单向2＋1车道龙门架计算）✧第20项需要另外根据需求进行开发。

✧上述的工程量清单，是按照主干道单向2＋1车道进行估算的，如果单方向超过2＋1个车道，设备数量需要进行调整。

✧以上设备不含工程实施部分，未罗列相关的工程辅材。

✧第13、14项为北京信路威公司自有产品，其他产品用户可以自购。