低功耗的集成栅极驱动电路设计
mos管栅极快速放电电路
mos管栅极快速放电电路摘要：1.MOS 管的特点和应用2.MOS 管栅极快速放电电路的原理3.MOS 管栅极快速放电电路的设计要点4.MOS 管栅极快速放电电路的应用实例5.总结正文：一、MOS 管的特点和应用MOS 管（金属- 氧化物- 半导体场效应晶体管）是一种常见的半导体器件，具有高输入阻抗、低噪声和低功耗等特点。
在电子电路中，MOS 管被广泛应用于放大器、开关和振荡器等电路。
在点焊机中，MOS 管常用于栅极驱动电路，以实现焊接过程中的精确控制。
二、MOS 管栅极快速放电电路的原理在点焊机中，MOS 管栅极快速放电电路的作用是使MOS 管在焊接过程中能够快速关断，以实现焊接电流的精确控制。
当焊接电流达到设定值时，栅极快速放电电路将使MOS 管栅极电荷迅速放电，从而降低MOS 管的导通状态，使焊接电流停止流动。
三、MOS 管栅极快速放电电路的设计要点在设计MOS 管栅极快速放电电路时，需要考虑以下要点：1.选择合适的MOS 管型号：根据焊接电流的大小和焊接过程的要求，选择合适的MOS 管型号。
2.确定栅极电阻值：为了使MOS 管栅极电荷能够快速放电，需要选择合适的栅极电阻值。
通常情况下，栅极电阻值应选取在10-100kΩ之间。
3.选择合适的放电二极管：放电二极管的选择应根据MOS 管的额定电压和放电电流来确定，以保证在焊接过程中能够快速放电。
4.确保电路的稳定性：在设计电路时，应注意确保电路的稳定性，避免因电路参数不合理而导致的MOS 管损坏或焊接效果不佳。
四、MOS 管栅极快速放电电路的应用实例在点焊机中，MOS 管栅极快速放电电路可以实现焊接电流的精确控制，提高焊接质量。
例如，在汽车制造行业中，点焊机常用于汽车车身的焊接，MOS 管栅极快速放电电路可以实现焊接电流的精确控制，以确保焊接质量达到汽车制造行业的标准。
五、总结MOS 管栅极快速放电电路在点焊机中起着关键作用，可以实现焊接电流的精确控制，提高焊接质量。
mos管栅极驱动电阻
MOS管（金属氧化物半导体场效应晶体管）是一种广泛应用在电子电路中的开关元件。
它的栅极驱动电阻是影响其工作性能的重要因素之一。
栅极驱动电阻的选择主要取决于MOS管的输入电容和驱动电压。
一般来说，栅极驱动电阻越大，MOS管的开关速度越慢，但是可以减小栅极驱动电流，降低功耗。
反之，如果栅极驱动电阻过小，可能会导致栅极驱动电流过大，从而引起MOS管的误导通或损坏。
在实际电路设计中，通常需要根据具体的应用需求来选择合适的栅极驱动电阻。
例如，对于高速开关应用，可能需要选择较小的栅极驱动电阻以加快开关速度；而对于低功耗应用，可能需要选择较大的栅极驱动电阻以降低功耗。
此外，还需要注意的是，由于MOS管的输入电容会随着温度的变化而变化，因此在设计栅极驱动电路时，还需要考虑到温度的影响。
一般来说，随着温度的升高，MOS管的输入电容会增大，因此需要相应地增大栅极驱动电阻。
总的来说，栅极驱动电阻的选择是一个需要综合考虑多种因素的问题，包括MOS管的特性、电路的应用需求以及环境条件等。
只有正确地选择了栅极驱动电阻，才能确保MOS管的正常工作和良好的性能。
在实际电路设计中，工程师通常会使用一些专业的电路设计软件来进行栅极驱动电阻的选择和优化。
这些软件可以根据用户输入的参数和条件，自动计算出最优的栅极驱动电阻值，大大提高了设计效率和准确性。
栅极驱动芯片分类
栅极驱动芯片分类
栅极驱动芯片是一种电子元器件，广泛应用于电路中。
它们可以帮助控制晶体管或场
效应晶体管的栅极电压，从而实现电路的稳定工作。
在栅极驱动芯片的分类方面，市场上
主要有以下几种类型：
1. 单通道栅极驱动芯片
单通道栅极驱动芯片只能控制一个晶体管或场效应晶体管的栅极电压，只有一个通道。
这种芯片通常用于低功耗、低应变和小信号的应用中。
通常，单通道栅极驱动芯片具有低
功耗和高可靠性等优点。
多通道栅极驱动芯片可以控制多个晶体管或场效应晶体管的栅极电压，具有多个独立
的通道。
这种芯片通常用于高功率应用，如电机控制、电源管理和自动驾驶系统等。
其通
道数通常在 4 或 8 个以上，能够满足多路开关控制的需求。
相比于双通道栅极驱动芯片，多通道栅极驱动芯片具有更高的功率和更广泛的控制范围。
集成型栅极驱动芯片具有更强的集成度，通常包括控制逻辑、功率放大器和开关元件
等多个功能单元。
这种芯片通常用于高效率应用，如 LED 照明、电机控制和电源管理等。
其优点包括高度集成的能力、高效率和较小的 PCB 布局，能够有效提高整个设备的性能
和可靠性。
总的来说，栅极驱动芯片是电路控制中不可或缺的元器件，它们有不同的分类方式和
应用场景。
因此，在选择栅极驱动芯片时，应根据实际需求选择合适的类型，以达到更好
的性能和效果。
高速MOSFET栅极驱动电路的设计与应用指南
高速MOSFEMOSFET T栅极驱动电路的设计与应用指南摘要本文将展示一个用来设计高速开关应用所需的高性能栅极驱动电路的系统性方案。
它综合了各方面的信息，可一次性解决一些最常见的设计问题。
因此，各个层面的电力电子工程师都值得一读。
文中分析了一些最流行的电路方案及其性能，包括寄生元件、瞬间和极端工作条件的影响。
首先，文章对MOSFET技术和开关操作进行了大致讨论，从简单问题逐渐转向复杂问题，并详细讲述了低端和高端栅极驱动电路以及交流耦合和变压器隔离式方案的设计程序。
另外，文章还专门用一个章节的内容来讨论同步整流器应用中MOSFET的栅极驱动要求。
最后，本文还提供了多个分步骤的设计案例。
简介MOSFET，全称为金属氧化物半导体场效应晶体管，是电子产品领域各种高频高效开关应用的关键元器件。
FET技术发明于1930年，比双极晶体管还要早大约20年，这一点令人感到意外。
最早的信号级FET晶体管出现在20世纪50年代末，而功率MOSFET则是在70年代中期问世的。
如今，数百万的MOSFET 晶体管被集成到了各种电子元器件中，从微控制器到“离散式”功率晶体管。
本话题的重点在于各种开关模式电源转换应用中功率MOSFET的栅极驱动要求。
Design And Application GuideFor High Speed MOSFET Gate Drive CircuitsBy Laszlo BaloghABSTRACTThe main purpose of this paper is to demonstrate a systematic approach to design high performance gate drive circuits for high speed switching applications. It is an informative collection of topics offering a “one-stop-shopping” to solve the most common design challenges. Thus it should be of interest to power electronics engineers at all levels of experience.The most popular circuit solutions and their performance are analyzed, including the effect of parasitic components, transient and extreme operating conditions. The discussion builds from simple to more complex problems starting with an overview of MOSFET technology and switching operation. Design procedure for ground referenced and high side gate drive circuits, AC coupled and transformer isolated solutions are described in great details. A special chapter deals with the gate drive requirements of the MOSFETs in synchronous rectifier applications.Several, step-by-step numerical design examples complement the paper.INTRODUCTIONMOSFET – is an acronym for Metal Oxide Semiconductor Field Effect Transistor and it is the key component in high frequency, high efficiency switching applications across the electronics industry. It might be surprising, but FET technology was invented in 1930, some 20 years before the bipolar transistor. The first signal level FET transistors were built in the late 1950’s while power MOSFETs have been available from the mid 70’s. Today, millions of MOSFET transistors are integrated in modern electronic components, from microprocessors, through “discrete” power transistors.The focus of this topic is the gate drive requirements of the power MOSFET in various switch mode power conversion applications. MOSFET TECHNOLOGYThe bipolar and the MOSFET transistors exploit the same operating principle. Fundamentally, both type of transistors are charge controlled devices which means that their output current is proportional to the charge established in the semiconductor by the control electrode. When these devices are used as switches, both must be driven from a low impedance source capable of sourcing and sinking sufficient current to provide for fast insertion and extraction of the controlling charge. From this point of view, the MOSFETs have to be driven just as “hard” during turn-on and turn-off as a bipolar transistor to achieve comparable switching speeds. Theoretically, the switching speeds of the bipolar and MOSFET devices are close to identical, determined by the time required for the charge carriers to travel across the semiconductor region. Typical values in power devices are approximately 20 to 200 picoseconds depending on the size of the device. The popularity and proliferation of MOSFET technology for digital and power applications is driven by two of their major advantages over the bipolar junction transistors. One of these benefits is the ease of use of the MOSFET devices in high frequency switching applications. The MOSFET transistors are simpler to drive because their control electrode is isolated from the current conducting silicon, therefore a continuous ON current is not required. Once the MOSFET transistors are turned-on, their drive current is practically zero. Also, the controlling charge and accordingly the storage time in the MOSFET transistors is greatly reduced. This basically1eliminates the design trade-off between on state voltage drop – which is inversely proportional to excess control charge – and turn-off time. As a result, MOSFET technology promises to use much simpler and more efficient drive circuits with significant economic benefits compared to bipolar devices.Furthermore, it is important to highlight especially for power applications, that MOSFETs have a resistive nature. The voltage drop across the drain source terminals of a MOSFET is a linear function of the current flowing in the semiconductor. This linear relationship is characterized by the R DS(on) of the MOSFET and known as the on-resistance. On-resistance is constant for a given gate-to-source voltage and temperature of the device. As opposed to the -2.2mV/°C temperature coefficient of a p-n junction, the MOSFETs exhibit a positive temperature coefficient of approximately 0.7%/°C to 1%/°C. This positive temperature coefficient of the MOSFET makes it an ideal candidate for parallel operation in higher power applications where using a single device would not be practical or possible. Due to the positive TC of the channel resistance, parallel connected MOSFETs tend to share the current evenly among themselves. This current sharing works automatically in MOSFETs since the positive TC acts as a slow negative feedback system. The device carrying a higher current will heat up more – don’t forget that the drain to source voltages are equal – and the higher temperature will increase its R DS(on) value. The increasing resistance will cause the current to decrease, therefore the temperature to drop. Eventually, an equilibrium is reached where the parallel connected devices carry similar current levels. Initial tolerance in R DS(on) values and different junction to ambient thermal resistances can cause significant – up to 30% – error in current distribution.Device typesAlmost all manufacturers have got their unique twist on how to manufacture the best power MOSFETs, but all of these devices on the market can be categorized into three basic device types. These are illustrated in Figure 1.Figure 1. Power MOSFET device types Double-diffused MOS transistors were introduced in the 1970’s for power applications and evolved continuously during the years. Using polycrystalline silicon gate structures and self-aligning processes, higher density integration and rapid reduction in capacitances became possible. The next significant advancement was offered by the V-groove or trench technology to further increase cell density in power MOSFET devices. The better performance and denser integration don’t come free however, as trench MOS devices are more difficult to manufacture.The third device type to be mentioned here is the lateral power MOSFETs. This device type is constrained in voltage and current rating due to its inefficient utilization of the chip geometry. Nevertheless, they can provide significant benefits in low voltage applications, like in microprocessor power supplies or as synchronous rectifiers in isolated converters.2The lateral power MOSFETs have significantly lower capacitances, therefore they can switch much faster and they require much less gate drive power.MOSFET ModelsThere are numerous models available to illustrate how the MOSFET works, nevertheless finding the right representation might be difficult. Mostof the MOSFET manufacturers provide Spice and/or Saber models for their devices, but these models say very little about the application traps designers have to face in practice. They provide even fewer clues how to solve the most common design challenges.A really useful MOSFET model which would describe all important properties of the device from an application point of view would be very complicated. On the other hand, very simple and meaningful models can be derived of the MOSFET transistor if we limit the applicabilityof the model to certain problem areas.The first model in Figure 2 is based on the actual structure of the MOSFET device and can be used mainly for DC analysis. The MOSFET symbol in Figure 2a represents the channel resistance and the JFET corresponds to the resistance of the epitaxial layer. The length, thus the resistance of the epi layer is a function of the voltage rating of the device as high voltage MOSFETs require thicker epitaxial layer.Figure 2b can be used very effectively to model the dv/dt induced breakdown characteristic of a MOSFET. It shows both main breakdown mechanisms, namely the dv/dt induced turn-on of the parasitic bipolar transistor - present in all power MOSFETs - and the dv/dt induced turn-onof the channel as a function of the gate terminating impedance. Modern power MOSFETs are practically immune to dv/dt triggering of the parasitic npn transistor due to manufacturing improvements to reduce the resistance between the base and emitter regions.It must be mentioned also that the parasitic bipolar transistor plays another important role. Its base – collector junction is the famous body diode of the MOSFET.Figure 2. Power MOSFET models34Figure 2c is the switching model of the MOSFET. The most important parasitic components influencing switching performance are shown in this model. Their respective roles will be discussed in the next chapter which is dedicated to the switching procedure of the device.MOSFET Critical ParametersWhen switch mode operation of the MOSFET is considered, the goal is to switch between the lowest and highest resistance states of the device in the shortest possible time. Since the practical switching times of the MOSFETs (~10ns to 60ns) is at least two to three orders of magnitude longer than the theoretical switching time (~50ps to 200ps), it seems important to understand the discrepancy. Referring back to the MOSFET models in Figure 2, note that all models include three capacitors connected between the three terminals of the device. Ultimately, the switching performance of the MOSFET transistor is determined by how quickly the voltages can be changed across these capacitors.Therefore, in high speed switching applications, the most important parameters are the parasitic capacitances of the device. Two of these capacitors, the C GS and C GD capacitors correspond to the actual geometry of the device while the C DS capacitor is the capacitance of the base collector diode of the parasitic bipolar transistor (body diode).The C GS capacitor is formed by the overlap of the source and channel region by the gate electrode. Its value is defined by the actual geometry of the regions and stays constant (linear) under different operating conditions.The C GD capacitor is the result of two effects. Part of it is the overlap of the JFET region and the gate electrode in addition to the capacitance of the depletion region which is non-linear. The equivalent C GD capacitance is a function of the drain source voltage of the device approximated by the following formula:DS1GD,0GD V K 1C C ⋅+≈The C DS capacitor is also non-linear since it is the junction capacitance of the body diode. Its voltage dependence can be described as:DS 2DS,0DS V K C C ⋅≈Unfortunately, non of the above mentioned capacitance values are defined directly in the transistor data sheets. Their values are given indirectly by the C ISS , C RSS , and C OSS capacitor values and must be calculated as: RSSOSS DS RSS ISS GS RSSGD C C C C C C C C −=−== Further complication is caused by the C GD capacitor in switching applications because it is placed in the feedback path between the input and output of the device. Accordingly, its effective value in switching applications can be much larger depending on the drain source voltage of the MOSFET. This phenomenon is called the “Miller” effect and it can be expressed as:()GD L fs eqv GD,C R g 1C ⋅⋅+=Since the C GD and C DS capacitors are voltage dependent, the data sheet numbers are valid only at the test conditions listed. The relevant average capacitances for a certain application have to be calculated based on the required charge to establish the actual voltage change across the capacitors. For most power MOSFETs the following approximations can be useful: offDS,spec DS,spec OSS,ave OSS,off DS,spec DS,spec RSS,ave GD,V V C 2C V V C 2C ⋅⋅=⋅⋅=The next important parameter to mention is the gate mesh resistance, R G,I . This parasitic resistance describes the resistance associated by the gate signal distribution within the device. Its importance is very significant in high speed switching applications because it is in between the driver and the input capacitor of the device, directly impeding the switching times and the5dv/dt immunity of the MOSFET. This effect is recognized in the industry, where real high speed devices like RF MOSFET transistors use metal gate electrodes instead of the higher resistance polysilicon gate mesh for gate signal distribution. The R G,I resistance is not specified in the data sheets, but in certain applications it can be a very important characteristic of the device. In the back of this paper, Appendix A4 shows a typical measurement setup to determine the internal gate resistor value with an impedance bridge.Obviously, the gate threshold voltage is also a critical characteristic. It is important to note that the data sheet V TH value is defined at 25°C and at a very low current, typically at 250μA. Therefore, it is not equal to the Miller plateau region of the commonly known gate switching waveform. Another rarely mentioned fact about V TH is its approximately –7mV/°C temperature coefficient. It has particular significance in gate drive circuits designed for logic level MOSFET where V TH is already low under the usual test conditions. Since MOSFETs usually operate at elevated temperatures, proper gate drive design must account for the lower V TH when turn-off time, and dv/dt immunity is calculated as shown in Appendix A and F.The transconductance of the MOSFET is its small signal gain in the linear region of its operation. It is important to point out that every time the MOSFET is turned-on or turned-off, it must go through its linear operating mode where the current is determined by the gate-to-source voltage. The transconductance, g fs , is the small signal relationship between drain current and gate-to-source voltage:GSD fs dV dI g =Accordingly, the maximum current of the MOSFET in the linear region is given by: ()fs th GS D g V V I ⋅−=Rearranging this equation for V GS yields the approximate value of the Miller plateau as a function of the drain current.fs D th Miller GS,g IV V +=Other important parameters like the source inductance (L S ) and drain inductance (L D ) exhibit significant restrictions in switching performance. Typical L S and L D values are listed in the data sheets, and they are mainly dependant on the package type of the transistor. Their effects can be investigated together with the external parasitic components usually associated with layout and with accompanying external circuit elements like leakage inductance, a current sense resistor, etc.For completeness, the external series gate resistor and the MOSFET driver’s output impedance must be mentioned as determining factors in high performance gate drive designs as they have a profound effect on switching speeds and consequently on switching losses.SWITCHING APPLICATIONSNow, that all the players are identified, let’s investigate the actual switching behavior of the MOSFET transistors. To gain a better understanding of the fundamental procedure, the parasitic inductances of the circuit will be neglected. Later their respective effects on the basic operation will be analyzed individually. Furthermore, the following descriptions relate to clamped inductive switching because most MOSFET transistors and high speed gate drive circuits used in switch mode power supplies work in that operating mode.Figure 3. Simplified clamped inductive switchingmodelThe simplest model of clamped inductive switching is shown in Figure 3, where the DC current source represents the inductor. Its current can be considered constant during the short switching interval. The diode provides a path for the current during the off time of the MOSFET and clamps the drain terminal of the device to the output voltage symbolized by the battery.Turn-On procedureThe turn-on event of the MOSFET transistor can be divided into four intervals as depicted in Figure 4.Figure 4. MOSFET turn-on time intervalsIn the first step the input capacitance of the device is charged from 0V to V TH. During this interval most of the gate current is charging the C GS capacitor. A small current is flowing through the C GD capacitor too. As the voltage increases at the gate terminal and the C GD capacitor’s voltage has to be slightly reduced. This period is called the turn-on delay, because both the drain current and the drain voltage of the device remain unchanged.Once the gate is charged to the threshold level, the MOSFET is ready to carry current. In the second interval the gate is rising from V TH to the Miller plateau level, V GS,Miller. This is the linear operation of the device when current is proportional to the gate voltage. On the gate side, current is flowing into the C GS and C GD capacitors just like in the first time interval and the V GS voltage is increasing. On the output side of the device, the drain current is increasing, while the drain-to-source voltage stays at the previous level (V DS,OFF). This can be understood looking at the schematic in Figure 3. Until all the current is transferred into the MOSFET and the diode is turned-off completely to be able to block reverse voltage across its pn junction, the drain voltage must stay at the output voltage level. Entering into the third period of the turn-on procedure the gate is already charged to the sufficient voltage (V GS,Miller) to carry the entire load current and the rectifier diode is turned off. That now allows the drain voltage to fall. While the drain voltage falls across the device, the gate-to-source voltage stays steady. This is the Miller plateau region in the gate voltage waveform. All the gate current available from the driver is diverted to discharge the C GD capacitor to facilitate the rapid voltage change across the drain-to-source terminals. The drain current of the device stays constant since it is now limited by the external circuitry, i.e. the DC current source.The last step of the turn-on is to fully enhance the conducting channel of the MOSFET by applying a higher gate drive voltage. The final amplitude of V GS determines the ultimate on-resistance of the device during its on-time. Therefore, in this fourth interval, V GS is increased from V GS,Miller to its final value, V DRV. This is accomplished by charging the C GS and C GD capacitors, thus gate current is now split between the two components. While these capacitors are being charged, the drain current is still constant, and the drain-to-source voltage is slightly decreasing as the on-resistance of the device is being reduced.6Turn-Off procedureThe description of the turn-off procedure for the MOSFET transistor is basically back tracking the turn-on steps from the previous section. Start with V GS being equal to V DRV and the current in the device is the full load current represented by I DC in Figure 3. The drain-to-source voltage is being defined by I DC and the R DS(on) of the MOSFET. The four turn-off steps are shown in Figure 5. for completeness.Figure 5. MOSFET turn-off time intervals The first time interval is the turn-off delay which is required to discharge the C ISS capacitance from its initial value to the Miller plateau level. During this time the gate current is supplied by the C ISS capacitor itself and it is flowing through the C GS and C GD capacitors of the MOSFET. The drain voltage of the device is slightly increasing as the overdrive voltage is diminishing. The current in the drain is unchanged.In the second period, the drain-to-source voltage of the MOSFET rises from I D⋅R DS(on) to the final V DS(off) level, where it is clamped to the output voltage by the rectifier diode according to the simplified schematic of Figure 3. During this time period – which corresponds to the Miller plateau in the gate voltage waveform - the gate current is strictly the charging current of the C GDcapacitor because the gate-to-source voltage is constant. This current is provided by the bypass capacitor of the power stage and it is subtracted from the drain current. The total drain current still equals the load current, i.e. the inductor current represented by the DC current source in Figure 3.The beginning of the third time interval is signified by the turn-on of the diode, thus providing an alternative route to the load current.The gate voltage resumes falling from V GS,Miller to V TH. The majority of the gate current is coming out of the C GS capacitor, because the C GDcapacitor is virtually fully charged from the previous time interval. The MOSFET is in linear operation and the declining gate-to-source voltage causes the drain current to decrease and reach near zero by the end of this interval.Meanwhile the drain voltage is steady at V DS(off)due to the forward biased rectifier diode.The last step of the turn-off procedure is to fully discharge the input capacitors of the device. V GSis further reduced until it reaches 0V. The bigger portion of the gate current, similarly to the third turn-off time interval, supplied by the C GScapacitor. The drain current and the drain voltage in the device are unchanged.Summarizing the results, it can be concluded that the MOSFET transistor can be switched between its highest and lowest impedance states (either turn-on or turn-off) in four time intervals. The lengths of all four time intervals are a function of the parasitic capacitance values, the required voltage change across them and the available gate drive current. This emphasizes the importance of the proper component selection and optimum gate drive design for high speed, high frequency switching applications.7Characteristic numbers for turn-on, turn-off delays, rise and fall times of the MOSFET switching waveforms are listed in the transistor data sheets. Unfortunately, these numbers correspond to the specific test conditions and to resistive load, making the comparison of different manufacturers’ products difficult. Also, switching performance in practical applications with clamped inductive load is significantly different from the numbers given in the data sheets.Power lossesThe switching action in the MOSFET transistorin power applications will result in some unavoidable losses, which can be divided into two categories.The simpler of the two loss mechanisms is the gate drive loss of the device. As described before, turning-on or off the MOSFET involves chargingor discharging the C ISS capacitor. When the voltage across a capacitor is changing, a certain amount of charge has to be transferred. The amount of charge required to change the gate voltage between 0V and the actual gate drive voltage V DRV, is characterized by the typical gate charge vs. gate-to-source voltage curve in the MOSFET datasheet. An example is shown in Figure 6.Figure 6. Typical gate charge vs. gate-to-sourcevoltage This graph gives a relatively accurate worst case estimate of the gate charge as a function of the gate drive voltage. The parameter used to generate the individual curves is the drain-to-source off state voltage of the device. V DS(off) influences the Miller charge – the area below the flat portion of the curves – thus also, the total gate charge required in a switching cycle. Once the total gate charge is obtained from Figure 6, the gate charge losses can be calculated as:DRVGDRVGATEfQVP⋅⋅=where V DRV is the amplitude of the gate drive waveform and f DRV is the gate drive frequency – which is in most cases equal to the switching frequency. It is interesting to notice that the Q G⋅f DRV term in the previous equation gives the average bias current required to drive the gate. The power lost to drive the gate of the MOSFET transistor is dissipated in the gate drive circuitry. Referring back to Figures 4 and 5, the dissipating components can be identified as the combination of the series ohmic impedances in the gate drive path. In every switching cycle the required gate charge has to pass through the driver output impedances, the external gate resistor, and the internal gate mesh resistance. As it turns out, the power dissipation is independent of how quickly the charge is delivered through the resistors. Using the resistor designators from Figures 4 and 5, the driver power dissipation can be expressed as:OFFDRV,ONDRV,DRVIG,GATELODRVGDRVLOOFFDRV,IG,GATEHIDRVGDRVHIONDRV,PPPRRRfQVR21PRRRfQVR21P+=++⋅⋅⋅⋅=++⋅⋅⋅⋅=In the above equations, the gate drive circuit is represented by a resistive output impedance and this assumption is valid for MOS based gate drivers. When bipolar transistors are utilized in the gate drive circuit, the output impedance becomes non-linear and the equations do not yield the correct answers. It is safe to assume that with low value gate resistors (<5Ω) most gate drive losses are dissipated in the driver. If R GATE is sufficiently large to limit I G below the output89current capability of the bipolar driver, the majority of the gate drive power loss is then dissipated in R GATE .In addition to the gate drive power loss, the transistors accrue switching losses in the traditional sense due to high current and high voltage being present in the device simultaneously for a short period. In order to ensure the least amount of switching losses, the duration of this time interval must be minimized. Looking at the turn-on and turn-off procedures of the MOSFET, this condition is limited to intervals 2 and 3 of the switching transitions in both turn-on and turn-off operation. These time intervals correspond to the linear operation of the device when the gate voltage is between V TH and V GS,Miller , causing changes in the current of the device and to the Miller plateau region when the drain voltage goes through its switching transition.This is a very important realization to properly design high speed gate drive circuits. It highlights the fact that the most important characteristic of the gate driver is its source-sink current capability around the Miller plateau voltage level. Peak current capability, which is measured at full V DRV across the driver’s output impedance, has very little relevance to the actual switching performance of the MOSFET. What really determines the switching times of the device is the gate drive current capability when the gate-to-source voltage, i.e. the output of the driver is at ~5V (~2.5V for logic level MOSFETs).A crude estimate of the MOSFET switching losses can be calculated using simplified linear approximations of the gate drive current, drain current and drain voltage waveforms during periods 2 and 3 of the switching transitions. First the gate drive currents must be determined for the second and third time intervals respectively:()G.I GATE HI MillerGS,DRV G3G.IGATE HI TH Miller GS,DRVG2R R R V V I R R R V V 0.5V I ++−=+++⋅−=Assuming that I G2 charges the input capacitor of the device from V TH to V GS,Miller and I G3 is the discharge current of the C RSS capacitor while the drain voltage changes from V DS(off) to 0V, the approximate switching times are given as:G3offDS,RSS G2THMillerGS,ISS I V C t3I V V C t2⋅=−⋅=During t2 the drain voltage is V DS(off) and the current is ramping from 0A to the load current, I L while in t3 time interval the drain voltage is falling from V DS(off) to near 0V. Again, using linear approximations of the waveforms, the power loss components for the respective time intervals can be estimated:Loff DS,Loff DS,I 2V T t3P32I V T t2P2⋅⋅=⋅⋅=where T is the switching period. The total switching loss is the sum of the two loss components, which yields the following simplifed expression:Even though the switching transitions are well understood, calculating the exact switching losses is almost impossible. The reason is the effect of the parasitic inductive components which will significantly alter the current and voltage waveforms, as well as the switching times during the switching procedures. Taking into account the effect of the different source and drain inductances of a real circuit would result in second order differential equations to describe the actual waveforms of the circuit. Since the variables, including gate threshold voltage, MOSFET capacitor values, driver output impedances, etc. have a very wide tolerance, the above described linear approximation seems to be a reasonable enough compromise to estimate switching losses in the MOSFET.Effects of parasitic componentsThe most profound effect on switching performance is exhibited by the source inductance. There are two sources for parasitic source inductance in a typical circuit, the sourceTt3t22I V P L DS(off)SW +⋅⋅=。
栅极驱动光耦
栅极驱动光耦
栅极驱动光耦是一种利用光耦合器来驱动MOSFET和IGBT的一种电路。
其主要原理是将普通电路中的桥式晶体管设备改造成一种完整的电路，并将其输出结果转换成光信号，再将光信号进行转换，最终作为栅极驱动电路驱动MOSFET和IGBT门极。
由于栅极驱动光耦能有效地隔离电路，因此它通常用于进行电气控制，如空调温度控制，可调整电压控制，智能电动汽车和其他运动控制等。
栅极驱动光耦的主要优点包括高分辨率、超低功耗、快速响应时间、可靠性高等。
栅极驱动光耦的结构主要由LED灯、电晕变压器、光敏电阻、栅极控制开关和桥式晶体管等组成，LED灯是输入电源给栅极控制开关，电晕变压器可以把输入压转换为低压电流，光敏电阻可以把LED灯输出的光转换成电流，栅极控制开关可以根据电流的大小来调整MOSFET 和IGBT的开关状态，最后，桥式晶体管可以把栅极的开关控制转化为实际的蜂鸣器控制。
此外，栅极驱动光耦还能够支持多种电压和电流，并且由于它具有较小的封装，具有较高的可靠性和抗干扰能力，因此比传统的桥式晶体管更有优势。
栅极驱动光耦可以通过外部控制设备，如电脑，来实现设备自动控制，从而提高了设备的使用效率。
总之，栅极驱动光耦具有多种优势，例如高分辨率、超低功耗、快速响应时间、可靠性高等，使它从传统的桥式晶体管中脱颖而出，是应用于智能电动汽车或其他运动控制系统的理想产品。
它能够减少
系统的设计复杂度，大大简化了系统的控制，并且能够满足各种复杂的工业应用。
隔离式栅极驱动电源模块
隔离式栅极驱动电源模块隔离式栅极驱动电源模块是一种常见的电子元件，广泛应用于各种电路中。
它的作用是提供稳定的电源信号，以驱动栅极，从而控制MOSFET等器件的导通和截止。
在本文中，将详细介绍隔离式栅极驱动电源模块的原理、特点以及应用。
一、原理隔离式栅极驱动电源模块主要由输入端、输出端和隔离电路组成。
输入端接受外部信号，经过隔离电路进行电气隔离，然后输出到输出端，用于驱动栅极。
隔离电路一般采用光耦隔离技术或者变压器隔离技术，能够有效地隔离输入端和输出端，避免信号干扰和传播。
二、特点1. 高隔离性：隔离式栅极驱动电源模块能够有效地隔离输入端和输出端，提供高度的信号隔离，避免信号干扰和传播。
2. 低功耗：隔离式栅极驱动电源模块采用高效的电路设计和低功耗元件，能够提供稳定的输出信号，同时降低功耗。
3. 高稳定性：隔离式栅极驱动电源模块采用稳定的电源电路和可靠的元件，能够在各种环境条件下保持稳定的输出信号。
4. 宽输入电压范围：隔离式栅极驱动电源模块具有宽输入电压范围的特点，能够适应不同电压输入的需求。
5. 多种保护功能：隔离式栅极驱动电源模块通常具有过压保护、过流保护、短路保护等多种保护功能，能够提高系统的可靠性和安全性。
三、应用隔离式栅极驱动电源模块广泛应用于各种电子设备和系统中。
主要应用领域包括：1. 电力系统：隔离式栅极驱动电源模块可以用于电力系统中的开关电源、逆变器、UPS等设备，提供稳定的驱动信号，实现电能的转换和传输。
2. 工业自动化：隔离式栅极驱动电源模块可以用于工业自动化系统中的PLC、变频器、伺服驱动器等设备，提供稳定的驱动信号，实现工业生产的自动化控制。
3. 电动汽车：隔离式栅极驱动电源模块可以用于电动汽车中的电机控制器、充电桩等设备，提供稳定的驱动信号，实现电动汽车的高效运行。
4. 新能源领域：隔离式栅极驱动电源模块可以用于太阳能发电系统、风力发电系统等新能源领域的电力控制设备，提供稳定的驱动信号，实现新能源的利用与管理。
IGBT驱动电路设计
IGBT驱动电路设计————————————————————————————————作者：————————————————————————————————日期：一种IGBT驱动电路的设计IGBT的概念是20世纪80年代初期提出的。
IGBT具有复杂的集成结构，它的工作频率可以远高于双极晶体管。
IGBT已经成为功率半导体器件的主流。
在10～100 kHz的中高压大电流的范围内得到广泛应用。
IGBT进一步简化了功率器件的驱动电路和减小驱动功率。
1 IGBT的工作特性。
IGBT的开通和关断是由栅极电压来控制的。
当栅极施以正电压时，MOSFET内形成沟道，并为PNP晶体管提供基极电流，从而使IGBT导通。
此时从N+区注入到N-区的空穴（少子）对N-区进行电导调制，减小Ⅳ区的电阻R dr ，使阻断电压高的IGBT也具有低的通态压降。
当栅极上施以负电压时。
MOSFET内的沟道消失，PNP晶体管的基极电流被切断，IGBT即被关断。
在IGBT导通之后。
若将栅极电压突然降至零，则沟道消失，通过沟道的电子电流为零，使集电极电流有所下降，但由于N-区中注入了大量的电子和空穴对，因而集电极电流不会马上为零，而出现一个拖尾时间。
2 驱动电路的设计2.1 IGBT器件型号选择1）IGBT承受的正反向峰值电压考虑到2-2.5倍的安全系数，可选IGBT的电压为1 200 V。
2）IGBT导通时承受的峰值电流。
额定电流按380 V供电电压、额定功率30 kVA容量算。
选用的IGBT型号为SEMIKRON公司的SKM400GA128D。
2.2 IGBT驱动电路的设计要求对于大功率IGBT，选择驱动电路基于以下的参数要求：器件关断偏置、门极电荷、耐固性和电源情况等。
门极电路的正偏压VGE负偏压-VGE和门极电阻RG的大小，对IGBT的通态压降、开关时间、开关损耗、承受短路能力以及dv／dt电流等参数有不同程度的影响。
门极驱动条件与器件特性的关系见表1。
基于upf的低功耗设计方法研究与实现
摘要摘要随着集成电路设计技术的不断发展及半导体工艺的进步，芯片的集成度、复杂度不断提高并且工作频率也得到大幅度提升，这导致芯片的功率密度显著增大，其工作时产生的功耗急剧增加。
功耗的增加增大了芯片测试的难度，同时对芯片的散热和封装提出了更加严苛的要求。
另外，为了符合节能规范的要求以及迫于市场的压力，降低芯片的功耗已是大势所趋。
功耗已成为VLSI设计优化中继速度、面积之后另一个须考虑的重要因素。
本课题来源于实习期间所做的项目，研究了集成电路的低功耗设计方法并对显卡芯片中的一个接口模块进行了低功耗设计。
本文首先研究了集成电路中功耗的组成（包括静态功耗与动态功耗）和各种低功耗设计方法。
其次，本文还研究了统一功率格式UPF标准以及用UPF进行低功耗设计的流程。
通过把功耗相关信息统一描述在一个UPF文件中，并在整个集成电路设计流程中都采用这个UPF文件所提供的功耗意图，从而在很大程度上降低了低功耗设计的复杂度以及风险。
然后采用以下低功耗设计技术对接口模块进行低功耗设计：1)多阈值电压技术：用多阈值电压库进行综合，即采用one-pass流程。
2)门控时钟技术：在逻辑综合阶段，利用工具Design Compiler自动完成时钟门控单元的插入，无需修改RTL代码。
3)多电压域和门控电源技术：在逻辑功能描述正确的基础上，使用UPF来描述低功耗设计的意图、指标及参数，并且完成了基于UPF的逻辑综合。
最后，利用形式验证工具Formality对原始的RTL+ UPF文件与综合后的门级网表+新产生的UPF文件UPF’进行了等价性检查，用工具VCS对RTL和UPF进行了带电源信息的仿真（Power Aware Simulation），以此验证了低功耗设计的正确性。
此外，在低功耗设计正确的基础上，本文通过分析和比较采用不同技术进行低功耗设计前后的功耗结果，得出了以下结论：采用多阈值电压技术后明显改善了静态功耗，门控时钟技术可显著降低动态功耗，而门控电源和多电压域技术能同时降低动态功耗和静态功耗，可最大程度地节省功耗。
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低功耗的集成栅极驱动电路设计廖聪维，陈韬，郑灿，张盛东北京大学，深圳研究生院，深圳，518055摘要：本文分析了传统集成栅极驱动电路的功耗与TFT尺寸、工作温度和存储电容的关系，发现集成栅极驱动电路的功耗与存储电容关系密切。
在对传统集成栅极驱动电路低功耗分析的基础上，本文介绍了一种基于多相交叠时钟的集成栅极驱动电路。
这种新电路具有低功耗的优点，且有效栅极驱动时间延长。
尤其在存储电容小，高温场合，这种集成栅极驱动电路节约功耗超过60%。
关键词：非晶硅（a-Si：H），薄膜晶体管（TFT），栅极驱动器1 引言在TFT LCD产业界，集成栅极驱动电路(GIA, gate driver in array)引起了广泛的兴趣[1]。
这是因为它具有减少外围IC数量及其连线、降低显示模组成本、提高显示面板分辨率和弯折度等优点[2]。
然而，显示器的总体功耗会由于栅极驱动器的集成而增加。
尤其是移动显示中，低功耗设计对于延长电池使用时间、增强移动设备的续航能力非常必要。
因此，集成栅极驱动电路的低功耗设计显得非常重要。
已有研究采用多相时钟的设计，降低时钟跳变频率减少集成栅极驱动电路的功耗[3]。
但是，很少有研究成果根据集成栅极驱动电路的特点做专门的低功耗优化设计。
本文研究了传统集成栅极驱动电路的功耗与器件参数的关系，发现存储电容值与功耗密切相关。
进而报道了一种新的低功耗集成栅极驱动电路。
2 电路分析图1是一种针对WXGA(800*1280) TFT LCD应用的传统的栅极驱动单元电路。
其由四部分构成：预充电部分、自基金项目：深圳市重点实验室提升项目(CXB201005260065A)Email: zhangsd@ 举上拉部分、下拉部分和低电平维持部分[4]。
其中，预充电部分包括T1; 自举上拉部分包括T2; 下拉部分为T3和T4；低电平维持部分包括C1，C2，T5~T8。
低电平维持部分的作用是抑制时钟馈通效应、防止电路的内部节点和输出节点上噪声电荷积累。
集成栅极驱动电路功耗包括静态功耗和动态功耗，以动态功耗为主。
集成栅极驱动电路的功耗表达式：()2CCK H L CKP V V f≈-,C CK, V H/V L和f CK分别是时钟信号负载电容，时钟信号的高/低电平，时钟信号的频率。
从功耗表达式可以看出，减少功耗的措施包括：(1) 减少电压幅度；(2)降低时钟频率；(3)减少负载电容等。
但是，减少电压幅度会导致TFT的驱动能力不足，输出信号的延迟时间增加。
降低时钟频率则容易增加电路的复杂程度。
同时，负载电容受限于工艺而不容易减少。
V I1图1 传统栅极驱动单元电路Fig. 1. a conventional a-Si:H TFT gate driver schematic很少有研究者讨论过C1对栅极驱动电路性能的影响。
图2 是C1分别为1，3和5 pF 时，功耗(800级)随温度的变化。
C1的增加能够显著地减少功耗。
P o w e r (W )TEMP图2功耗随温度的变化(C1分别为1，3和5 pF)。
Fig. 2. evolution of power consumption for different temperature with C1 of 1, 3, 5 pF.C G (p F )V G (V)图3 TFT 栅极电容与栅极电压的关系 Fig. 3. C G versus gate voltage for TFT.功耗的值之所以受C1的影响很大，是因为低电平维持阶段，V Q 的跳变幅度与T2的栅-漏电容和C1之间的分压有关，这就是所谓“时钟馈通”效应。
另一方面， T2的栅-漏电容又受到V Q 跳变电压的调制。
图3是栅极电容C G (包括栅-源电容、栅-漏电容)与栅极电压的关系。
若T2管工作于负栅压区或者V T 以上区域， C G 是一个与V G 无关的常量；但是在V T 附近，C G 随V G 有较大的变化。
在低电平维持阶段，T2管由于时钟馈通效应短暂地开启。
由于低电平维持电路，T2的短暂开启并不会造成输出节点上电荷的积累。
但是，T2管的短暂开启足以造成T2的栅极电容非线性地增加。
因此，增加C1能够抑制T2的开启程度，从而减少功耗。
但是，电路的面积也会因为C1的增加而变大。
因此，有必要改进电路结构，在快速度、低功耗、减少电路面积等方面实现优化。
3 新电路的分析和讨论测试所用的a-Si:H TFT 以及栅极驱动电路均在Gen 4.5 工艺线上完成了加工。
所制作的a-Si:H TFT 采用了背沟道刻蚀型的结构，沟道长度为4.5 μm ，栅极-源漏电极交叠宽度为2 μm 。
根据Agilent B1500A 实际测试得到的a-Si:H TFT 特性，利用业界广泛认可的RPI 模型(Level 35)对TFT 进行模拟[5], [6]。
所采用的模拟工具是SmartSpice 。
V V A V V CV DV I2V OV I3(b)-2002040-10010-10010V Q (V )V O (V )200100V D (V )Time ( s)(c)图 4 一种新的栅极驱动单元电路(a)电路图; （b ）时序图； (c)模拟输出波形图Fig. 4. the proposed gate driver circuit (a) schematic of one unit circuit; (b) waveforms; (c) simulated waveforms of V Q , V O and V D .图4示意了一种新的栅极驱动单元电路(a)电路图; （b ）时序图；(c)模拟输出波形图。
这种电路是利用时钟信号的交叠，在V A 变成高电平之前将驱动管T2的栅极稳定在低电平，从而抑制时钟馈通效应、降低功耗。
这种新的栅极驱动电路的工作分为以下几个阶段：(1) 预充电阶段(t1)在t1阶段，V I1为高电平，V Q 通过T1和T4被充电到高电平。
当V Q 大于V T2时，T2在V A 变成高电平之前被打开。
此时由于V A 为低电平，V O 为低电平。
(2) V O 上拉阶段(t2)在t2阶段，V A 变为高电平，V O 通过提前开启的T2被满幅度地上拉到高电平，V Q 被自举到高电平。
在t2阶段的前半段，虽然V D 和V I1都是高电平，但是V Q 被自举到比V D 和V I1更高电位状态，因此V Q 的自举不会受到影响。
(3) V O 下拉阶段(t3)在t3阶段T3是关断的。
这是因为V I2和V I3都为高电平。
Q 点处于悬浮状态，这是因为与Q 点相连的T1，T4和T5都为关断。
所以，T2仍然是开启的。
V A 变为低电平，所以V O 被下拉到低电平。
从而，T2不仅完成了V O 的上拉，而且完成了V O 的下拉。
(4) Q 放电阶段(t4)在t4阶段，T3被打开。
这是因为V I3为高电平，而V I2变为低电平。
从而Q 点被下拉到低电平。
在V A 再次变成高电平之前，T2被关断。
(5) 低电平维持阶段(t5)在t5阶段，T7和T8分别在V A 和V C 控制下轮流导通。
当V A 跳变为高电平时，T7的栅极通过C2被耦合到为高电平，因此T7的栅-源电压大于V T7，从而T7被打开。
在V C 跳变为高电平时，T8的栅-源电压大于V T8，从而T8被打开。
从而V O 由于T7和T8的轮流导通而保持为低电平。
P o w e r (W )C1 (pF)图5 传统栅极驱动电路和新电路的功耗与C1关系，温度为20, 70O CFig. 5. power consumption comparison of conventional and the proposed gate driver versus C1 for temperature of 20, 70O C.由V D 控制的T4在V A 跳变为高电平之前而打开。
因此在t5阶段，Q 点总能够先于V A 的高电平都来而被连接到低电平。
这对于减小时钟馈通对Q 点电压跳变的影响很有效。
另一方面，当V A 为高电平时，T5也会打开，Q 点连接到V O,这对于抑制时钟馈通效应也有很好的作用。
图 4 (c)示意了新的栅极驱动单元电路的模拟波形图。
比较图3和图6(c)发现，新电路的Q点的电压更平稳。
因此，新电路的时钟信号负载电容更小。
图5 是分别在20和70O C时，传统、新集成栅极驱动电路的功耗与电容C1的关系。
如第二部分所述，必须采用较大的C1才能够减少传统电路的功耗。
而新电路的功耗值与C1几乎无关，且新电路的功耗值小于传统电路的功耗值。
尤其在较高温度下，功耗的减少量超过60%。
4 总结本文分析了传统集成栅极驱动电路的功耗来源，首次报道了存储电容对于传统集成栅极驱动电路的功耗影响较大。
本文介绍了一种基于多相时钟的集成栅极驱动电路，其具有较低的功耗。
这种新的集成栅极驱动电路有望应用于高性能的移动显示应用场合。
参考文献[1] J. H. Oh, J. H. Hur, Y. D. Son, K. M. Kim, S. H. Kim, E. H. Kim, J. W. Choi, S. M. Hong, J. O. Kim, B. S. Bae, and J. Jang, “2.0 inch a-Si:H TFT-LCD with Low Noise Integrated Gate Driver,” in Proc. S ID Symp. Dig., 2005, pp. 942-945.[2] S. H. Moon, Y. S. Lee, M. C. Lee, B. H. Berkeley, N. D. Kim and S. S. Kim, “Integrated a-Si:H TFT Gate Driver Circuits on Large Area TFT-LCDs,” in Proc. SID Symp. Dig., 2007, pp. 1478-1481.[3] I. Hwang, S. Moh, M. Lee, and E. Lee, “Design of Integrated a-Si Gate Driver Circuits for Low Power Consumption,” in Proc. SID Symp. Dig., 2008, pp. 842-845.[4] J. W. Choi, M. S. Kwon, J. H. Koo, J. H. Park, S. H. Kim, D. H. Oh, S. Lee, and J. Jang, "Noble a-Si:H Gate Driver with High Stability," in Proc. SID Symp. Dig., 2008, pp. 1227-1230.[5] J. W. Choi, J. I. Kim, S. H. Kim, and J. Jang, "Highly Reliable Amorphous Silicon Gate Driver Using Stable Center-Offset Thin-Film Transistors," IEEE Trans. Electron Devices, vol. 57, no. 9, pp. 2330-2334, 2010.[6] C. L. Lin, C. D. Tu, M. C. Chuang, and J. S. Yu, "Design of Bidirectional and Highly Stable Integrated Hydrogenated Amorphous Silicon Gate Driver Circuits," IEEE J. Display Technology, vol. 7, no.1, pp. 10-18, 2011.Low Power Integrated a-Si:H Gate Driver DesignCongwei Liao, Tao Chen, Can Zheng, Shengdong ZhangShenzhen graduate school, Peking University, Shenzhen, 518055Email: zhangsd@AbstractPower consumption of a conventional integrated a-Si:H gate driver scheme is analyzed by exploring the relationship of power consumption with the TFTs’size, operation temperature, and storage capacitor. For the first time, it is found power consumption is in close relationship with the storage capacitor. A new integrated a-Si:H gate driver based on multi-phase clocks is proposed, which owns the merits of low power consumption, extended effective scanning time. Especially at high operation temperature and small storage capacitor, the proposed gate driver saves more than 60% power consumption compared with the conventional one.Key Words: a-Si:H, TFT, gate driver。